Introduction
Nowadays, antennas are rapidly developing, the characteristics of antennas, such as high gain, circularly polarization (CP), and simple structure are getting more and more attention. Due to the Fabry–Perot resonator antenna (FPRA) simple structure, easy integration, and high gain, it is a nice method for design [Reference Li, Liu, Jia and Guo1–Reference Xie and Wang10]. Today, there are several methods to realize the circularly polarized FPRA (CP-FPRA): CP feed antenna and polarization conversion metasurface (PCM).
In papers [Reference Li, Liu, Jia and Guo1, Reference Zarbakhs, Akbari, Samadi and Sebak2], proposed sequential feed antenna arrays are used as CP feed antennas for FPRA. Both antennas achieve good CP radiation but have complex feed networks. The feed antennas are reconfigurable CP antennas [Reference Ji, Qin, Guo, Ding, Fu and Gong3, Reference Zhang, Liu, Jia and Zhang4]. They can operate in different polarization states, but they have no advantages in processing and maintenance. Papers [Reference Orr, Goussetis and Fusco5–Reference Xie and Wang10] show a suitable PCM to achieve CP radiation. Two orthogonal linear polarization (LP) waves present a 90° transmission phase difference to form a CP wave. For CP-FPRA, the PCM is usually a metasurface with reflective and transmissive properties. The conventional receiver-transmitter (RT) metasurfaces have a metallized plane to separate the lower and upper layers of receiver and transmitter patches [Reference Lau and Hum11, Reference Pan, Huang, Ma, Jiang and Luo12]. And incident waves by metallized through-holes achieve coupling [Reference Kaouach, Dussopt, Lanteri, Koleck and Sauleau13–Reference Diaby, Clemente, Pham, Sauleau and Dussopt15]. When receives the LP and radiates the CP, the PCM shows a performance of polarization conversion. Therefore, it is feasible to design the CP-FPRA with a PCM based on RT unit.
In this paper, a novel RT-based PCM unit with two metallized via-holes is proposed for the design of CP-FPRA. Metallized via-holes are used to connect the receiver and transmitter of the PCM unit and correct the current to achieve LP to CP conversion. Meanwhile, it maintains high reflectivity to ensure a high gain of the CP-FPRA. PCM and the feed antenna are combined to form the CP-FPRA. The proposed CP-FPRA achieves high gain and wide axial ratio (AR). The 3 dB AR bandwidth of 8.6% (22.3–24.3 GHz) and a peek gain reaches 14.2 dBic at 24 GHz.
Antenna design and analysis
CP-FPRA configuration
The configuration of the proposed CP FPRA is shown in Figure 1. The antenna consists of a PCM and a feed antenna. The h represents the distance between PCM and the feed antenna, it determines the final profile of the antenna. The LP waves are radiated from the feeder and reflected in the cavity (formed by the PCM and the ground plane). At the same time, the LP resonance is realized in the cavity, and the PCM convert the LP waves in the cavity into CP waves. The resonance generation of CP-FPRA and the conversion of LP to CP are spatially separated.
Geometry of the proposed CP-FP antenna. (a) CP-FPRA oversize. (b) CP-FPRA side view.
Operating principles
CP-FPRA can work when Equation (1) is satisfied, h (distance between the PCM and the feed antenna) determines the final profile of the antenna [Reference Trentini16].
\begin{equation}{\varphi _S} - {\varphi _D} - \frac{{4\pi h}}{{{\lambda _0}}} = 2N\pi ,N = 0, \pm 1, \pm 2,\ldots\end{equation} where 
${\lambda _0} = \frac{c}{{{f_0}}}$, 
${f_0}$is the operating frequency, 
${\varphi _S}$is the reflection phases of the PCM, and 
${\varphi _D}$is the reflection phases of ground plane.
According to the full-wave analysis and measurement verification, the calculated relationship between the directivity D of CP-FPRA and the reflection coefficient 
$\Gamma $of PCM can be obtained as follows (Equation (2)) [Reference Foroozesh and Shafai17]:
\begin{equation}D = 10{\log _{10}}\frac{{1 + |\Gamma |}}{{1 - |\Gamma |}}\end{equation} It found the directivity D of the CP-FPRA is higher when the PCM reflection coefficient 
$\Gamma $is larger, thus enhancing the antenna gain. Therefore, the high reflectivity of the PCM ensures a high gain of the CP-FPRA. In addition, the following conditions are required to realize the polarization conversion: the magnitude of the transmission coefficients is 
${T_{xx}}$= 
${T_{yx}}$and the transmission phase difference is (
$\varphi {T_{xx}} - \varphi {T_{yx}}$) = 90° [Reference Xie, Wang, Li, Liang and Gao8].
Design and analysis of feed antenna
A CP-FPRA with good performance requires a good feed antenna. The feed antenna as shown in Figure 1(a), the two layer dielectric substrates are made of Rogers 5880 (εr = 2.2, tanδ = 0.0009), the thickness h 0 = 0.508 mm. The top layer consists of four identical square patches as the radiation patch, the metallized plane is printed between the two dielectric substrates. And the bottom layer is fed using a microstrip line and transmits energy through an L-shaped probe.
Figure 2 shows the simulated current distribution of the feed antenna on one T (T is the period of the excitation signal) at 24 GHz. At t = 0, the current is mainly concentrated on the square patches, and the current flows along the negative x-axis. At t = T/4, the current is mainly concentrated at the middle rectangular patch, and the current direction is along the positive x-axis. When t = T/2 and 3T/4, the current distribution is similar to at t = 0 and T/4, but has the opposite direction. Therefore, the feed antenna periodically radiates x-polarization.
The feed antenna surface current distribution of the top patch at 24 GHz. (a) t = 0, (b) t = T/4, (c) t = T/2, (d) t = 3T/4.
Design process of PCM
Development of PCM unit
Figure 3 shows configuration of the original unit cell. The period of unit p = 6 mm (
$0.48{\lambda _0}$), it has two same substrates (“Rogers 5880,” 
${\varepsilon _{\text{r}}}{\text{ = }}2.2$, 
${h_1}$= 1.57 mm) and three layers of metal patches. The top layer is a cross-shaped structure (as the transmitter), the middle layer is a metallized plane, and the bottom layer is a square patch (ws = 4.9 mm), acts as the receiver. A metallized via is set in the x-axis offset from the origin 
${x_0}$(
${x_0}$= 0.8 mm).
Configuration of the original unit cell. (a) 3D sketch. (b) Detailed structure.
Figure 4 is developed from Figure 3 has two differences. First, a rectangle is sequentially added at the end of the cross-patch on the upper layer. This addition can increase the electrical length of the patch, improve the transmission performance of the unit, and ensure the stability at a certain angle (maintaining consistent performance when the electromagnetic wave is incident at various angles). Second, added a metallized via in the y-axis to correct the current and achieve better polarization conversion. Figure 4(b) shows the simulation model of the proposed PCM unit in the High Frequency Structure Simulator (HFSS). Port 1 and Port 2 are set Floquet ports. And Port 1 is the excitation port.
Configuration of the proposed polarization conversion unit. (a) Detailed structure. (b) Simulation model of the unit.
Figure 5 provides data to support the above. Figure 5(a) shows simulated S parameters of the original unit cell with metallized via in the x-axis. The reflectivity of co-polarization reaches 0.98, but the reflectivity of cross-polarization is almost 0. The 
${T_{xx}}$is keep above 0.2, while the 
${T_{yy}}$and 
${T_{yx}}$are almost 0. So, the unit can transmit a part of the x-polarization and prevent the y-polarization. And it cannot realize the polarization conversion. Correspondingly, Figure 5(b) shows the metallized via in the y-axis. The unit can pass through a part of the y-polarization and prevent the x-polarization, and still not have the polarization conversion function.
(a) Simulated S parameters of the original unit cell with metallized via in the x-axis. (b) Simulated S parameters of the original unit cell with metallized via in the y-axis. (c) Simulated S parameters of the original unit cell with metallized vias in the x-axis and y-axis.
Figure 5(c) illustrates the reflection and transmission of the unit with two equally spaced metallized vias. The second metallized via unaffected the reflection performance, and the co-polarization transmission of the unit. But can improve the transmission of cross-polarization of the unit. So, a suitable 
${x_0}$can achieve 
${T_{xx}}$= 
${T_{yx}}$.
Figure 6 shows the variation of the transmission coefficients with the position 
${x_0}$. When 
${x_0}$= 1.4 mm, the transmission coefficient 
${T_{xx}}$of the proposed unit is equal to 
${T_{yx}}$at 24 GHz. And the transmission performance is improved compared with original cell. The transmission phase difference decreases as the 
${x_0}$increases. When 
${x_0}$= 1.4 mm, the phase difference is 90°, at 24 GHz. This satisfies the conditions for a polarization conversion.
Effect of 
${x_0}$on the S parameters of the proposed polarization conversion unit cell. (a) Transmission coefficients. (b) Phase difference.
Figure 7 shows the variation of the transmission and reflection coefficients with the p (period of unit). When p = 6 mm (0.48
${\lambda _0}$), the transmission coefficient 
${T_{xx}}$= 
${T_{yx}}$, at 24 GHz. And the reflection magnitude first increases and then decreases, as the value of p increases. From Equation (2), a larger the reflection magnitude results in a higher antenna gain. So, p = 6 mm (0.48
${\lambda _0}$) was chosen, it is good for enhancing antenna radiation.
Effect of p on the S parameters of the proposed polarization conversion unit cell. (a) Transmission coefficients. (b) Reflection coefficients.
Properties of PCM unit
Figure 8 presents the transmission and reflection characteristics of the optimized PCM unit with normal incidence. The transmission coefficients 
${T_{xx}}$= 
${T_{yx}}$ and has a phase difference of 90° at 24 GHz. The reflection phase of the unit is −120° at 24 GHz. After optimization with HFSS, h is set to 6.9 mm. It is determined the PCM consists of 5 × 5 units with dimensions of 30 × 30 mm2.
Simulated S parameters of the proposed unit cell versus frequency. (a) Transmission coefficients. (b) Reflection coefficients.
To demonstrate the angular stability of the proposed unit. Figure 9 plots the S parameters of the unit cell versus the incident angle at 24 GHz. The xoz, yoz, and phi = 
$45^\circ $planes are simulated. When the incident angle is less than 
$15^\circ $, horizontal polarization transmission coefficient is equal to vertical polarization transmission coefficient in any plane. Meanwhile a phase difference between 
$90^\circ \pm 15^\circ $, it can achieve polarization conversion while maintaining a high co-polarization reflectivity.
Simulated S parameters of the proposed unit cell versus the angle. (a) 24 GHz in the xoz plane. (b) 24 GHz in the yoz plane. (c) 24 GHz in the phi = 45° plane.
Current analysis of PCM
To further explore the polarization conversion function of the proposed unit, the current distribution of the proposed units at 24 GHz is analyzed. Since the feed antenna radiates x-polarization, each unit is excited by x-polarized waves.
Figure 10 illustrates the current distribution of metallized vias in the proposed unit at 24 GHz. When t = 0, the current of metallized via-holes in the x-direction is upward, and t = 3 T/4, the current of metallized via-holes in the y-direction flows upward. Correspondingly, at t = T/4, the current of metallized via-holes in the x-direction is the same as the current of metallized via-holes in the y-direction at t = 0. So, compared with the current of metallized via-holes in the x-direction, the current of metallized via-holes in the y-direction is delayed by T/4 time. In other word, the unit has a 90° phase difference.
The current distribution of metallized via in the proposed polarization conversion unit.
Figure 11 shows the current distribution of the upper radiation patch at 24 GHz. The current alternates in quarter cycles with clockwise rotation, enabling Left-handed circular polarization (LHCP) radiation. Therefore, the proposed unit has a polarization conversion function.
The current distribution of the upper patch of the proposed polarization conversion unit. (a) t = 0, (b) t = T/2, (c) t = 3T/4, (d) and t = T.
Array analysis and experimental results
To further explain the impact of grating lobes on antenna radiation. Figure 12 shows the effect of PCM with different values of p on the normalized radiation performance of CP-FPRA. When p = 5 mm (0.4
${\lambda _0}$) and p = 7.5 mm (0.6
${\lambda _0}$), the levels of cross-polarization are excessively high, and the effect of the gating lobes is noticeable. p = 6 mm (0.48
${\lambda _0}$) significantly enhances radiation performance and effectively suppresses the effects of gating lobes.
Simulated normalized radiation patterns of the antenna with change p. (a) xoz plane at 24 GHz. (b) yoz plane at 24 GHz.
Figure 13(a) shows a fabricated prototype of the proposed CP-FPRA. Figure 13(b) shows the simulated and measured results of |
${S_{11}}$|. The simulated |
${S_{11}}$| < −10 dB of CP-FPRA is 12.3% (22.8–25.8 GHz), the measured result reaches 12% (22.83–25.7 GHz). The simulated |
${S_{11}}$| < −10 dB of feed antenna is 9.4% (23.3–25.6 GHz). It found the antenna impedance bandwidth is increased by adding PCM. Figure 13(c) plots the gain of CP-FPRA and the feed antenna, AR of CP-FPRA. The top dotted arrow points to the gain and the bottom dotted arrow points to the AR. The antenna achieves peak gain at 24 GHz with a simulated result of 14.41 dBic and a measured result of 14.2 dBic. A gain enhancement of 5.5 dBi (from 8.7 dBi to 14.2 dBic) is achieved by adding the PCM. The simulated 3 dB AR band of the CP-FPRA is 9% (22.2–24.3 GHz), and the measured result is 8.6% (22.3–24.3 GHz). The difference between the test results and the simulated results may be in antenna assembly and testing procedures.
(a) Photographs of fabricated antenna. (b) 
${S_{11}}$of simulated and measured results. (c) AR and gain of simulated and measured results.
Figure 14 shows the simulated and measured normalized radiation patterns of the CP-FPRA. At 24 GHz, the cross-polarization levels are lower than −34 dB, shows low side lobe (<−18 dB), and radiates LHCP. The 3 dB half-power beamwidth extends up to ±11° in the xoz and yoz planes. At 24.3 GHz (the frequency band edge), the cross-polarization levels are higher. However, the main beam is still LHCP. The measured and simulated results almost match.
Simulated and measured normalized radiation patterns of the antenna. (a) xoz plane at 23.7 GHz. (b) yoz plane at 23.7 GHz. (c) xoz plane at 24 GHz. (d) yoz plane at 24 GHz. (e) xoz plane at 24.3 GHz. (f) yoz plane at 24.3 GHz.
Table 1 presents a comparison between the proposed antenna and other antennas in previous articles. Compared with Refs [Reference Xie, Wang, Li, Liang and Gao8, Reference Yang, Yang and Li18] and [Reference Wang and Zhang19] its AR bandwidth and array size have a significant advantage. Compared with the Refs [Reference Zhang, Liu, Jia and Zhang4] and [Reference Liu, Zhang, Yin and Lu20], the antenna achieves the conversion of LP to CP, while significantly improving the gain. The proposed CP-FPRA is capable of CP radiation by utilizing a PCM with a feed antenna positioned normally, while enhancing the feed antenna gain with a compact profile.
Comparison of the proposed antenna with previously reported antennas
Abbreviations: L: linear, C: circular, PM: polarization mode, BW: bandwidth, PG: peak gain, GE: gain enhancement, NME: number of metasurface elements.
Conclusion
In this paper, a novel PCM unit is proposed. The metallized vias can achieve polarization conversion and maintain stable transmission and reflection coefficients. The CP-FPRA is obtained by combining this PCM and the feed antenna. The proposed CP-FPRA has high gain, wide AR, and a relatively low profile. It achieves a peak gain of 14.2 dBic at 24 GHz, compared to the feed antenna has a gain enhancement of 5.5 dBi (from 8.7 dBi to 14.2 dBic). The bandwidth of AR < 3 dB is 8.6% (22.3–24.3 GHz). The proposed design is a new method for achieving antenna high-gain and CP radiation.
Funding statement
This work was supported in part by the Shandong Natural Science Foundation of ZR2023MF038 and the National Natural Science Foundation of China (61701278).
Competing interests
The author declares that there is no conflict of interest regarding the publication of this paper.
Xin He was born in Dezhou, China, in 1999, and graduated from Jining University in 2022 with a B.S. degree in computer science and technology, and she is currently pursuing a M.S. degree in engineering at Qufu Normal University. Her research interests include Fabry–Perot resonant antenna and multifunctional metasurface.
Wenhua Qin Graduated from the Department of Physics, Qufu Normal University, in1992. Working mainly in the research areas of embedded system applications, Internet of things technology, intelligent information processing, and hadron physics.
Kaize Wang was born in Manzhouli, China, in 1999. He received his bachelor’s degree in electronic information engineering from Beijing University of Technology in 2021. He is currently pursuing a master’s degree in engineering at Qufu Normal University. His current research interests include 5G millimeter wave magnetoelectric dipole antennas and arrays.
Juan Xu was born in 1982 and graduated from Nanjing University of Science and Technology in 2016 with a Ph.D. in electronic science and technology. She is currently an associate professor of Cyber Science and Engineering, at Qufu Normal University. Her research interests mainly include microwave/millimeter wave circuits and systems, integrated circuits, antenna array optimization and beam forming, multifunctional electromagnetic metasurface, and terahertz biosensing.
