Antenna design and simulation

Our objective is to design a wideband MIMO antenna. The process started with designing of a single-antenna element, and after successive iterations, the structure of the single element antenna has been finalized. The successive design stages of the single-antenna element with their frequency responses have been depicted in Fig. 1.

Figure 1.

Successive design stages and responses of single-antenna element.

Design-I consists of a simple rectangular-shaped patch. The shape and size of the rectangular patch have been modified in Design II. The advantages of spade-shaped patches have been studied in [Reference Gong, Tong, Tian and Gao26–Reference Di and Li28]. The goal of radiating structures with spade shapes is to aid in the development of several resonant frequencies, which later combine to form a wide frequency band. So, with the aim of increasing the bandwidth, the patch is given the shape of a spade in Design III, which exhibits better response in comparison with the former designs (Fig. 1).

A MIMO antenna structure has many advantageous features such as higher data rate, time diversity, frequency diversity, reduced signal distortion, and higher accuracy. Common MIMO configurations are 2 × 2, 3 × 3, 4 × 4, 8 × 8 etc. In order to maintain the design simplicity and uniformity and space conservation and enhanced isolation between antenna elements, a 2 × 2 MIMO antenna structure has been proposed here.

The process started with the blueprint for Antenna I (Fig. 2), which consists of two spade-shaped radiators constructed of copper that have been annealed and mounted on top of an RT/Duroid substrate with a dielectric constant of 4.4 and a loss tangent of 0.0004.

Figure 2.

MIMO antenna with orthogonal spade-shaped radiators (Antenna I).

The radiators are kept orthogonal to improve the mutual isolation and to maintain the diversity in signal transmission/reception. A continuous ground plane made of copper material exists beneath the 1.0 mm thick substrate. Two orthogonally organized microstrip lines that are matched to a load impedance of 50 Ω for exciting the radiators. Using altered geometrical structures as radiating elements, it is possible to increase the impedance bandwidth and add new resonant frequencies, as described in [Reference Cho, Kim, Choi, Lee. and Park29]. After a certain number of simulation rounds, each radiating element has been adjusted to produce the most optimized response. The offset-feeding methodology suggested here offers a significantly larger impedance spectrum when compared to the traditional center feeding method [Reference Jan and Kao30]. The benefits of aligning the radiating elements perpendicularly include better isolation and polarization variability. The planar construction has the following dimensions: 20 × 20 × 1 mm³. The structure demonstrated a multiband frequency response covering three distinct frequency bands 20–21.5 GHz, 22–23 GHz, and 26–29 GHz after simulation in CST Microwave Studio Suite^TM. In this instance, the isolation level obtained is −24 dB (Fig. 3).

Figure 3.

S-parameter vs. frequency response of Antenna I.

It should be observed that at 20.3, 22.5, and 27 GHz, respectively, there are essentially three resonant modes. Figure 4 depicts the antenna surface current flow in Antenna I for three distinct resonating frequencies. Figure 5 shows the field magnitude at appropriate frequency points.

Figure 4.

Pattern of surface current flow of Antenna I with Port 1 in excited mode at: (a) 20.3 GHz, (b) 22.5 GHz, (c) 27 GHz.

Figure 5.

Far-field magnitude of Antenna I at: (a) 20.3 GHz, (b) 22.5 GHz, and (c) 27 GHz.

Deformation or a flaw in the antenna ground plane is a very effective technique that may be used to enhance frequency responsiveness. Defective ground structures aid in bandwidth control, impedance matching, and surface-wave suppression [Reference Rehman, Sheta and Alkanhal31]. A slotted structure in the shape of a rhombus produces a wide bandwidth, offers low tolerance for mass alignment, and is also simpler to fabricate or print. The installation of the rhombic slot in the ground plane, which is responsible for the formation of new resonant modes, can be shown to have increased the magnetic field’s strength as well as the E-field’s magnitude. The ground of Antenna I has a rhombic-shaped slot cutout of it with the goal of increasing bandwidth even more. Antenna II is the newly formed structure (Fig. 6). The dimensions of the rhombic slot have been chosen by fine tuning the slot dimensions namely slot width, Sw, and slot length, Sl. The best response is obtained when the slot dimensions are kept as Sw = 7.0 mm and Sl = 16.0 mm. The change of antenna frequency response with respect to variation in antenna parameters Sw and Sl has been depicted in Fig. 7.

Figure 6.

MIMO antenna with orthogonal spade-shaped radiators and rhombic slot in the ground plane (Antenna II).

Figure 7.

Change in antenna frequency response due to variation of Sw and Sl.

In simulation, Antenna II displayed a wide bandwidth, spanning 20–28 GHz, and a little improvement in isolation level of |S₂₁| = 25 dB (Fig. 8). It is evident that there is a deviation in the flow of antenna current along the ground plane as a result of the presence of the rhombic slot.

Figure 8.

S-parameter vs. frequency response of Antenna II.

It is clear that the addition of a ground plane slot has caused a deviation in the flow of antenna surface current, increasing the average magnetic field intensity and, consequently, expanding the impedance bandwidth. Figure 9 shows the surface current flow pattern of Antenna II with Port 1 activated at the resonance frequencies of 20.8 GHz, 22.2 GHz, 24.3 GHz, and 27 GHz. The field magnitude of Antenna II at the same frequency points is depicted in Fig. 10.

Figure 9.

Pattern of surface current flow of Antenna II with Port 1 in excited mode at: (a) 20.9 GHz, (b) 22.1 GHz, (c) 24.5 GHz, d) 27 GHz.

Figure 10.

Far-field magnitude of Antenna II at: (a) 20.9 GHz, (b) 22.1 GHz, (c) 24.5 GHz, and (d) 27 GHz.

The single-antenna element provides multiband response. A MIMO structure has been incorporated to introduce the advantageous features of the same such as higher data rate, time diversity, frequency diversity, reduced signal distortion, and higher accuracy. In order to maintain the design simplicity and uniformity and space conservation and enhanced isolation between antenna elements, two radiating structures, placed orthogonally have been placed.

There has been a negligible change in the frequency response of Antenna I due to the insertion of the second element. However, there has been an abrupt change in the response when there is a deformation in the ground plane. The reason for the same can be explained with the aid of antenna surface current distribution.

As observed from Fig. 4, the antenna surface current in the ground plane is distributed evenly. Defected ground structures, as discussed earlier, have the ability to induce resonant modes in the frequency response of the antenna, which mainly occurs due to the disturbance in the antenna surface current flow. With the insertion of the rhombic slot, the surface current is disturbed, which can be observed from Fig. 9. The resonant modes so generated thereby come closer to one another, which in turn results in the increase in the overall bandwidth.

Researchers working on antennas have taken advantage of parasitic strips to improve isolation, as seen in [Reference Ding, Gao, Qu and Yin32]. Parasitic components are crucial in lowering the mutual coupling between antenna elements. Here, we have added a diagonal T-shaped stub to Antenna III to isolate the radiating components (Fig. 11).

Figure 11.

MIMO antenna with orthogonal spade-shaped radiators, rhombic slot in the ground plane and T-shaped parasitic element (Antenna III).

Figure 12 shows the S-parameter vs frequency response of Antenna III. The surface current flow pattern of Antenna III with Port 1 excited at 20.6 GHz, 22.2 GHz, 24.2 GHz, and 27.1 GHz is depicted in Fig. 13. Figure 14 shows Antenna III’s field strength at the same frequency points. As seen in Fig. 13, the presence of a parasitic stub prevents the flow of antenna current across ports, serving as a de-coupler and enhancing isolation. The antenna current is clearly seen to remain concentrated around the parasitic strip. Therefore, it is conceivable that the parasitic stub that is positioned between the two radiators is the cause of the deviation in the antenna current’s path between ports.

Figure 12.

S-parameter vs frequency response of Antenna III.

Figure 13.

Pattern of surface current flow of Antenna III with Port 1 in excited mode at: (a) 20.8 GHz, (b) 22.4 GHz, (c) 24 GHz, and (d) 27.1 GHz.

Figure 14.

Far-field magnitude of Antenna III at: (a) 20.8 GHz, (b) 22.4 GHz, (c) 24 GHz, (d) 27.1 GHz.

The S-parameter vs. Freq. response charts for Antennas I, II, and III are shown in Figs. 15 and 16. Three distinct frequency bands (20–21.5 GHz), (22–23 GHz), and (26–29 GHz) have been acquired for Antenna I. In this instance, a maximum isolation level of −44 dB was attained. Antenna II displayed a broad bandwidth in simulation, spanning from 20.0 GHz to just over 28 GHz, despite the fact that the maximum isolation level (−45 dB) remained essentially same. Our suggested structure, Antenna III, on the other hand, produced a broad bandwidth of 20.0–28.1 GHz combined with an increased maximum isolation level of |S₂₁| = 47 dB, which is unquestionably a remarkable achievement in isolation improvement and reduced crosstalk as cross talk depends on the value of |S₂₁|. Higher the value of |S₂₁|, lesser will be the cross talk. The following are Antennas I, II, and III’s varied dimensions: W = 20.0, L = 20.0, S1 = 4.0, S2 = 16.0, Sw = 7.0, Sl = 16.0, Fw = 2.0, F1 = 4.25, F2 = 2.25, P1 = 4.0, P2 = 6.5, P3 = 6.5, P4 = 3.0, P5 = 4.75, P6 = 3.25, p = 0.75, l = 4.0, r = 0.75 and s = 0.75 mm.

Figure 15.

S₁₁ vs. frequency response of Antenna I, II, and III.

Figure 16.

S₂₂ vs.frequency response of Antenna I, II, and III.

Antenna fabrication, measurement, and analysis

With expert assistance, the antenna prototype is created (Fig. 17). The walls are covered with urethane foam absorbers.

Figure 17.

Fabricated prototype of Antenna III (front and rear views).

It may be mentioned here that as a reference antenna for measurement purposes, a standard gain pyramidal horn antenna (WR-22) made of Al-material with a nominal gain of 24.7 dBi has been employed. The Antenna under test is fixed to the positioning system, and N5-230A Vector Network Analyzer (VNA) was utilized to measure the antenna responses.

The entire operating and measurement process has been depicted in Fig. 18. The antenna under test (AUT) has been placed on the positioning system, which acts as the Receiver (Rx) antenna and the Transmitter (Tx) antenna used here is the horn antenna. Signal is radiated from the directional horn antenna and the same is received by the AUT. The signal responses are measured with the help of the VNA.

Figure 18.

Schematic showing antenna operating and measurement process.

The S-parameter vs. Freq. response of the antenna is shown in Fig. 19. The values of the S-parameters are well matched for simulation and measurement as well. As seen in Fig. 19, the antenna produced a wide impedance bandwidth of 21.5–28.5 GHz and an enhanced isolation level of |S₂₁| = 25 dB, which is unquestionably a significant advancement in isolation enhancement.

Figure 19.

S-parameter vs. frequency response of Antenna III (simulated vs. measured).

An antenna with a standard gain is kept on the turntable and calibrated in the bore-sight direction to measure the gain of the AUT. The antenna needs to be positioned correctly. The power level (relative) is now determined with the aid of VNA, which enables us to determine the relative gain of the antenna under inquiry [Reference Balanis33]. Gain being a function of directivity, we may determine the gain/radiation pattern by altering the gain in different directions. Figure 20 shows the AUT’s gain and radiation efficiency.

Figure 20.

Rad. Eff. and gain plot of Antenna III (proposed antenna).

Along the whole bandwidth, antenna gain varies from 7 to 9.3 dBi, while radiation efficiency varies from 65% to 85%. Envelope-Correlation-Coefficient or ECC, is crucial when examining a MIMO antenna’s diversity characteristic. It is a measure of the degree of independency of the antenna radiation patterns. Low value is an indication of higher isolation between multiple MIMO ports. Several methods can be used to calculate ECC. However, the “3D Radiation Pattern” approach is dependable and precise. Equation (1) may be used to compute the ECC for a MIMO antenna as,

(1)\begin{equation}{\rho _e} = \frac{{\iint_{4\pi } {}|{P_1}(\phi ,\varphi )_{}^*{P_2}(\phi ,\varphi )|_{}^2\,d\Omega }}{{\iint_{4\pi } {}|{P_1}(\phi ,\varphi )|_{}^2\,d\Omega \iint_{4\pi } {}|{P_2}(\phi ,\varphi )|_{}^2\,d\Omega }}\end{equation}

where, ${P_i}(\phi ,\varphi ) = P_\theta ^i(\phi ,\varphi )\overrightarrow {{a_\theta }} + P_\phi ^i(\phi ,\varphi )\overrightarrow {{a_\phi }} $ is the ith element’s radiation field [Reference Liu, Yang, Jia and Guo34]. Using Equation (1), the ECC for antennas may be determined using S-parameters [Reference Hallbjorner35] as follows:

(2)\begin{equation}{\rho _e}(\,p,q,R) = \frac{{|{C_{p,q}}(R){|^2}}}{{\prod\limits_{k = (p,q)} {[1 - {C_{k,k}}(R)]} }}\end{equation}

where ${C_{p,q}}(R)$is given as follows:

\begin{equation*}{C_{p,q}}(R) = \sum\limits_{n = 1}^R {S_{p,n}^*} S_{n,q}^2\end{equation*}

ECC can also be used to determine diversity gain (DG) using Equation (3) as follows:

(3)\begin{equation}DG = 10\sqrt {1 - {{({\rho _e})}^2}} \end{equation}

Diversity techniques are used to reduce the fading effect. Diversity gain increases the concentration of probability density function (PDF) of the instantaneous value of signal-to-noise ratio (SNR), around its average value [Reference Jing, Lin, Liang, Shen, Shen, Du, Zhang, Yang and Tang36]. Low ECC and high DG are characteristics of an effective MIMO diversity antenna [Reference Biswas, Pattanayak and Chakraborty37]. The suggested MIMO antenna’s ECC/DG is shown in Fig. 20 and was calculated using the radiation field approach and S-parametric techniques, respectively.

The proposed MIMO antenna, as seen in the image, has produced low ECC (0.45 dB) and high DG (9.3 dB), which are unquestionably encouraging numbers for strong antenna performance. Channel Capacity is the maximum rate of transfer of information through a channel. The Channel Capacity Loss, or CCL is also a crucial parameter to be accounted for. For optimal performance with light channel capacity deterioration and good information transformation, it should be 0.4 Bits/s/Hz. Equation (4) can be used to calculate CCL as follows:

(4)\begin{equation}CCL = - {\log _2}\det ({\gamma ^r})\end{equation}

where ${\gamma ^r}$ is the correlation matrix in this case [Reference Nej, Ghosh, Ahmad, Kumar, Ghaffar and Hussein38]. Figure 21 shows the proposed MIMO antenna’s CCL, which is around 0.37 Bits/s/Hz.

Figure 21.

ECC and DG plot of Antenna III (proposed antenna).

We can better comprehend the impact of fading by using Mean Effective Gain, or MEG. The ratio of the mean values of accepted to incident antenna power along a certain direction is known as MEG [Reference Chae, Kawk, Park and Lee39]. Equation (5) can be used to compute it as follows:

(5)\begin{equation}ME{G_i} = 0.5{\nu _{p,rad}} = 0.5(1 - {\sum\limits_{y - 1}^N {|{S_{pq}}|} ^2})\end{equation}

The MEG is less than 0.3 dB across the whole operational bandwidth, as can be seen in Fig. 22.

Figure 22.

MEG and CCL plot of Antenna III (proposed antenna).

Total active reflection coefficient, often known as TARC, is a crucial metric to assess diversity characteristics, system performance, and impedance bandwidth in addition to the S-parameters. TARC for the suggested antenna can be calculated using equation (6) and the S-parameters as follows:

(6)\begin{equation}\Gamma = \frac{{\sqrt {|{S_{pp}} + {S_{pq}}{e^{j\varphi }}{|^2} + |{S_{pq}} + {S_{qq}}{e^{j\varphi }}{|^2}} }}{{\sqrt 2 }}\end{equation}

Here, φ is the phase for activating the input port. Input and output ports’ reflection coefficients are represented by S_pp and S_qq, respectively, and the isolation between the two ports is represented by S_pq/S_qp [Reference Chae, Kawk, Park and Lee39]. The TARC has been assessed in relation to a few excitation phase angles. Figure 23 shows the TARC curves for 0°, 90°, and 180°, which show good behavior over the whole operational spectrum.

Figure 23.

TARC plot of Antenna III (proposed antenna).

A time domain response analysis is also required to determine the antenna group delay. Equation (7) yields

(7)\begin{equation}{\text{the Group Delay}},\;{\gamma _g} = \frac{{\Delta \alpha }}{{\Delta \beta }}\end{equation}

where Δα and Δβ are the transmitted signal’s phase deviation and frequency deviation [Reference Chae, Kawk, Park and Lee39], respectively. Group delay refers to the delay of the signal traveling through the test device as a function of frequency. It is an evaluation of the time required for the modulated signal to traverse the network [Reference Sharma, Tiwari, Singh and Kanaujia40, Reference Mahto, Singh, Sinha, Alibakhshikenari, Khan and Pau41]. The suggested MIMO antennas in this design are spaced apart by 30 cm in both the face-to-face and side-by-side orientations to analyze the signal latency. Signal distortion happens as a result of group delay variation. Figure 24 shows the characteristic of antenna group delay. The fluctuation is only 0.5 ns, as can be seen in the image, which is an acceptable margin.

Figure 24.

Group delay plot of Antenna III (sim. vs. mea.).

In this aspect, it may be mentioned that due to the emergence of independent platform communication system, there exists a probability of co-site interferences. Eminent researchers around the world have proposed several techniques for cancellation of this signal interference. Various adaptive co-site Interference Cancellation Systems (ACICS) have been proposed.

In this regard, effective techniques of broadband interference cancellation as in [Reference Jiang and Li42, Reference Jiang, Liu, Li, Zhao and Wu43] may be adopted to enhance the performance of the proposed MIMO antenna system. The interference cancellation bandwidth can be improved by delay matching, which has been validated after simulation followed by results of measurement.

In order to measure the radiation pattern across several planes (XZ/YZ), the AUT’s orientation with respect to the reference horn antenna must be modified. Figure 25 shows the 2D-polar plot, which includes the simulated and measured antenna radiation patterns (including Cross-Pol. and Co-Pol.) for the XZ and YZ planes at frequencies of 20.5 GHz, 24.3 GHz, and 27.2 GHz, respectively. The pattern has few nulls and is slightly omnidirectional. Well-isolated co and cross polarization patterns are a positive indicator of the practical applicability of the antenna under test.

Figure 25.

Radiation pattern of proposed antenna at YZ/XZ-plane: (a) 20.5 GHz, (b) 24.3 GHz, and (c) 27.2 GHz.

The proposed MIMO antenna uses multiple input and output streams efficiently for improved signal quality and diversity. By introducing multiple antennas in the transmitting and receiving ends, multipath fading reduction and channel capacity improvement can be achieved. For this purpose, the antenna should be compact in shape and size, portable, and uncorrelated to each other. Our proposed MIMO antenna structure satisfies all the aspects in this regard.

The main challenge faced during the entire process is taking into account of the various errors during the measurement of the antenna gain. A simplified error budget has been prepared for the measurement of antenna gain, which is shown in Table 2.

Table 2.

A simplified error budget for the measurement of antenna gain