Antenna geometry and its analysis

The proposed dual-port MIMO (Multi Input Multi Output)antenna consists of dual-patch antenna (Fig. 1a) with a MS absorber structure (Fig. 1b) in place of the ground plane. It has the potential to be an RFID reader since it can function better in normal antenna mode and also serve as an absorber to decrease scattering. This may effectively lower the number of RFID systems that are incorrectly read in multipath environments. Figure 1c presents the shape and specific physical properties of the suggested antenna. The patch as well as MS absorber is printed on FR4 substrate (ε_r = 4.4 and tanδ = 0.02) with a thickness of 1.6 mm [Reference Ghaneizadeh, Mafinezhad and Joodaki39].

Figure 1.

Geometrical layout of two-port MIMO antenna (a) front view of stair shaped slot loaded patch; (b) front view of metasurface absorber; (c) 3D layout of proposed antenna.

Figure 2 illustrates the fluctuation in S-parameter for both single- and dual-port antennas in the absence of any MS absorber loading. The following scenarios are examined: (a) a single-port patch antenna with no stair slot loading; (b) a single-port patch antenna with stair slot loading; (c) a dual-port patch antenna with stair slot loading that is oriented the same way; and (d) a dual-port patch antenna with stair slot loading that is oriented in a mirror manner. Figure 2 yields several noteworthy observations. The proposed antenna’s impedance matching is improved by the stair-shaped slot loading; the isolation level is improved by more than 25 dB due to the mirror orientation of the slot loading; the reflection coefficient features of the single- and dual-port antenna are nearly the same, which is necessary for a MIMO radiator; and (iv) the slot loading also shifts the resonant frequency to the lower side. A microstrip antenna’s slot loading supplies the increment in the patch’s actual current path. As a result, the antenna becomes compact and the resonant frequency drops [Reference Chen, Qing and Chung40].

Figure 2.

S-parameter variation with single- and dual-port antenna without any MS absorber.

The axial ratio fluctuation with and without a stair-shaped slot (with equal and unequal arm length) is depicted in Fig. 3. When a stair-shaped slot with uneven arm length is loaded, Fig. 3 shows that the axial ratio falls below 3 dB within the working region. The stair-shaped slot separates the E-field into two orthogonal components with roughly equal amplitude. The arm length variation produces the 90^° phase shift between the orthogonal field components. It is a prerequisite for the creation of circular polarization (CP) waves [Reference Nasimuddin, Chen and Qing41].

Figure 3.

Axial ratio variation with and without loading of stair-shaped slot.

Currently in development is a MS absorber unit cell that can reduce multipath reflection and imprecise RFID reading. A perfect MS absorber is one that balances external coupling and internal losses to prevent reradiation. By balancing the transmission and reflection power coefficients, this is accomplished. Since the structure’s backside is completely covered in copper, the transmission coefficient is 0, allowing the absorptivity (A) to be calculated by

(1)\begin{equation}A = 1 - {\left| {{S_{11}}} \right|^2}{\text{ }}\end{equation}

The inherent losses of resonant unit cells significantly restrict the overall achievable absorption. Effective dielectric losses with the right impedance matchings or the addition of internal losses through the use of resistive materials are two potential ways to further enhance the absorption properties. Changing the external coupling as an alternative to changing the internal losses is another way to regulate the absorption process. This statement can be understood by calculating the input impedance of MS near to resonance. It is given as follows [Reference Balanis42]:

(2)\begin{equation}{Z\raise0.7ex\hbox{${_{in}}$} \!\mathord{\left/ {\vphantom {{_{in}} {{Z_0}}}}\right.} \!\lower0.7ex\hbox{${{Z_0}}$}} = \frac{{{\raise0.7ex\hbox{$1$} \!\mathord{\left/ {\vphantom {1 {{Q_{ex}}}}}\right.} \!\lower0.7ex\hbox{${{Q_{ex}}}$}}}}{{j2\delta + {\raise0.7ex\hbox{$1$} \!\mathord{\left/ {\vphantom {1 {{Q_{in}}}}}\right.} \!\lower0.7ex\hbox{${{Q_{in}}}$}}}}{\text{ }}\end{equation}

where $\delta $ = (ω − ω0)/ω0 is the detuned degree, ω0 is the angular resonant frequency, Qin and Qex are the internal and external Q-factors, respectively, and Z₀ is the impedance in free space. It is easily deduced from (A) that a locus of Zin forms a circle on the Smith chart with the diameter of 2Qin/(Qin + Qex), assuming that Qin and Qex are constant at the resonance. Since the impedance circle passes the origin (zero reflection) at the resonance, a complete absorption happens at the critical coupling of Qin = Qex. Whereas Qex originates from the external coupling between the guided waves in the MS and the normal incident waves, Qin originates from the conductor and dielectric losses in practical MSs [Reference Mumby and Yuan43, Reference Zhao, Zhu, Dong, Lv, Li, Guan, Shi and Zhang44]. Reflection coefficient (|S₁₁|) in terms of internal losses ($\frac{1}{{{\tau _0}}}$) and external coupling ($\frac{1}{{{\tau _e}}}$) is given as follows [Reference Zhao, Zhu, Dong, Lv, Li, Guan, Shi and Zhang44]:

(3)\begin{equation}\left| {{S_{11}}} \right| = \frac{{\left( {\frac{1}{{{\tau _e}}}} \right) - \left( {\frac{1}{{{\tau _0}}}} \right)}}{{\left( {\frac{1}{{{\tau _e}}}} \right) + \left( {\frac{1}{{{\tau _0}}}} \right)}}{ }\end{equation}

For $\left( {\frac{1}{{{\tau _e}}}} \right) = \left( {\frac{1}{{{\tau _0}}}} \right)$, i.e. the resonant frequency, there is no reflection, meeting the critical coupling requirement. It should be mentioned that the resonator is said to be over coupled when $\left( {\frac{1}{{{\tau _e}}}} \right) \gt \left( {\frac{1}{{{\tau _0}}}} \right)$, and under coupled when $\left( {\frac{1}{{{\tau _e}}}} \right) \lt \left( {\frac{1}{{{\tau _0}}}} \right)$. The rate of power escape from the resonant structure (MS absorber at resonance) is higher in the overcoupled area than the rate of internal dissipation. When considering the undercoupled scenario, internal losses surpass power leakage from the structure in terms of quantity.

The current study aims to meet the critical coupling requirement irrespective of the unit-cell size by optimizing that method in the megahertz range. We want to show how a straightforward unit cell with good reflection properties may be tuned to be the ideal absorber for normal incident waves on a substrate of the same dimensions and kind. Perfect absorption was attained by altering the external connection, which was accomplished by appropriately slicing diagonal slits on the metal circular patch. The patch underwent a first diagonal incision to lengthen the current channel and lower the resonance frequency. It also caused the region to become undercoupled as opposed to overcoupled. To symmetrize the E distribution locally with regard to the incident wave polarization – which is concentrated along the y-axis at the unit cell edges – a second slot was incorporated into the construction. This situation is explained in detail in Fig. 4, which shows the E-field distribution in three different scenarios: (a) an unslotted patch; (b) a circular patch with a single diagonal slit; and (c) a circular patch with two diagonal slits.

Figure 4.

E-field distribution at the top surface of the unit-cell design stages for an 868 MHz normal incident plane wave (a) circular patch without any slit (b) circular patch with one diagonal slit (c) circular patch with two diagonal slits.

It is crucial to keep in mind that the Poynting theorem links the imaginary component of permeability and permittivity to the real part of the mean power as time passes. Therefore, one of the most important factors in achieving 100% absorption is the dielectric tanδ. The Poynting theorem states that the real components of the meta-surface absorber (MSA) permittivity (ε) and permeability (μ) are engaged with the resonance of the MSA along the incident wave’s path [Reference Bao, Ruvio and Ammann45, Reference Augustin, Rao and Denidni46]. This explains why the stored power in the near-field of MEH switches between reactive magnetic and electric fields with each time-harmonic field cycle. While these components are necessary to generate the MSA resonance, they may be disregarded in order to compute the actual portion of average power [Reference Bao, Ruvio and Ammann45, Reference Baggen47]. On the other hand, at the resonance frequency of MEH, the imaginary components of ε and μ operate as power consumption, and a real resistance may mimic the EM response [Reference Bao, Ruvio and Ammann45]. It should be noted that an equal quantity of magnetic and electric energy may be carried by an EM wave traveling in free space. This is because, prior to exchanging their energy with the energy of the meta-atom, these traveling waves are in a balanced condition at the free-space medium [Reference Baggen47]. In fact, this balance is maintainable, and in a perfect MS with no reflections, the majority of the incoming wave’s actual average power is used at the loads. In this instance, the latter was accomplished by managing exterior bonding on a low-loss unit-cell material that had a tanδ of just 0.0015.

To demonstrate how the loss tangent influences absorbent properties, Fig. 5 simulates the absorption variation of the suggested unit cell with change in tanδ. At unity magnitude, the behavior of the lossless unit cell is similar to that of a typical HIS reflector, and at tanδ = 0.0015, nearly total absorption was achieved. Figure 6 displays the fluctuation of the S-parameter with and without the MS absorber after the unit cell has been converted into an array and put on the bottom part of the radiator. Figure 6 presents that the properties of the reflection coefficient are about the same in both scenarios. In other words, the intrinsic characteristics of the antenna remain unaffected by the suggested MS absorber’s loading. In all scenarios, there is about the same isolation among the ports. The gain fluctuation with and without an MS absorber is exposed in Fig. 7. It can be perceived from Fig. 7 that the inclusion of an MS absorber increases the gain value to 2–3 dBi. Notably, |S11| can approach 0 over a substantial percentage of the proposed absorber, in contrast to iigh-impedance surface, where the |S₁₁| magnitude at the resonant frequency is 1. Thus, the absorber in combination with the normal patch antenna not only yields a noticeable gain boost but may also help enhance the fading situation of RFID schemes by decreasing the severe complex replication interference and crash difficulties.

Figure 5.

Absorptivity variation with change in loss tangent value.

Figure 6.

S-parameter variation of dual-port antenna in presence/absence of MS absorber.

Figure 7.

Gain variation of dual-port antenna in presence/absence of MS absorber.

Experimental verification and discussion

This stage involves testing the suggested antenna prototype to validate the optimum outcomes. Figure 8 shows pictures of the built prototype of the proposed antenna. It is made on FR-4 dielectric, which has a dielectric constant of 4.4. The two-port radiator is positioned beneath the MS absorber. The suggested antenna’s measured and simulated S-parameter is displayed in Fig. 9. It is tested with an E5071C vector network analyzer that is based on Keysight. It is verified by Fig. 9 that the simulated and observed S-parameters have a strong correlation. The suggested antenna has an isolation level of more than 25 dB and operates efficiently in the frequency range of 710–980 MHz.

Figure 8.

Pictures of fabricated prototype of designed radiator (a) top view (b) MS absorber (b) 3D view of designed radiator.

Figure 9.

Measured/simulated S-parameter of proposed antenna design.

The suggested antenna’s observed and predicted axial ratio fluctuation is displayed in Fig. 10. Dual linear pattern measuring is used to measure it [Reference Nakamura and Fukusako48]. It is evident from Fig. 10 that the simulated and observed axial ratios accord rather well. The CP waves in the range of 835–894 MHz are supported by the suggested radiator. The simulated and observed LHCP (Left handed Circular Polarization)and RHCP (Right handed Circular Polarization) far-field pattern at 0.86 GHz in the XZ plane is displayed in Fig. 11. It can be perceived from Fig. 11 shows that LHCP waves with port-1 and RHCP waves with port-2 are supported by the suggested radiator. This indicates that the polarization diversity idea is supported by the suggested antenna. The generated and measured patterns correspond rather well. The suggested antenna design’s gain change in the broadside direction is shown in Fig. 12. Two antenna approach is used to measure it [Reference Nakamura and Fukusako48]. It is verified from Fig. 12 that the working band’s maximum antenna gain is about 3.0 dBi.

Figure 10.

Measured/simulated axial ratio variation of proposed antenna.

Figure 11.

Measured/simulated RHCP and LHCP of proposed radiator at 0.86 GHz (a) port-1 (b) port-2.

Figure 12.

Measured/simulated gain variation of proposed antenna toward broadside direction.

The diversity parameters, or the diversity gain (DG) and envelope correlation coefficient (ECC) fluctuation for the planned multiport radiator, are shown in Fig. 13. The similarity content between ports 1 and 2 is expressed by ECC. It should be low (less than 0.2) for effective radiator design [Reference Ghaneizadeh, Joodaki, Börcsök, Golmakani and Mafinezhad49]. The gain of a multiport radiator in a fading environment is expressed as DG. Its value for an efficient antenna is around 10.0 dB [Reference Ghaneizadeh, Joodaki, Börcsök, Golmakani and Mafinezhad49]. It is verified from Fig. 13 that, within the operational band, the value of DG is around 10.0 dB and the value of ECC is less than 0.1. Table 1 is used to evaluate the intended radiator’s performance with other current antennas in terms of gain, axial ratio bandwidth, and antenna size. Form Table 1, it is clearly observed that the proposed antenna design first time proposed a two-port MIMO along with CP characteristics and loaded with metamaterial absorber for UHF RFID application. The proposed antenna is compact in size as well as provides the good gain value as compared to other radiation structure.

Figure 13.

Measured/simulated ECC and DG variation of proposed antenna.

Table 1.

Performance assessment of proposed dual-port printed antenna with other existing antenna on the basis of absolute/axial ratio bandwidth, gain and antenna size