HIS unit cell design

Figure 1 shows the unit cell of the proposed HIS. The 8 × 8 array HIS structure is designed over an FR4 substrate (ε _r = 4.4) with a thickness of 1.6 mm. For fabrication simplicity, the HIS cell is kept planar, i.e. without vias. To make it polarization dependent, a slot is cut in the x-direction in the unit cell such that the effective x- and y-dimensions are not the same. Due to this asymmetry, the HIS behaves differently for the incident wave of x- and y-polarization [Reference Sievenpiper, Zhang, Broas, Opolous and Yablonovitch3]. Above this surface, the proposed antenna is placed over a distance of 0.5 mm from the HIS array.

Fig. 1.

HIS unit cell.

Antenna design

Figure 2 shows the bow tie slotted patch antenna. The proposed antenna is designed using an FR4 substrate (= 4.4) with a thickness of 1.6 mm and loss tangent is 0.0025. The antenna utilizes CPW feed. The dimension of patch is 20.70 × 23.50. The separation between ground and microstrip feed is 0.30 mm and the width of the microstrip line is 1.75 mm. The substrate is 60 × 60 separation between ground and patch is 14.25 mm.

Fig. 2.

Bow tie slotted patch antenna.

HIS operation

HISs can be designed in many ways such as to perform different functions viz. it can be utilized to have in-phase reflection, to control the propagation of surface waves and even to have variable surface impedance depending upon the incident wave polarization. These structures are periodic in nature and to maintain their periodicity their dimensions are less than the quarter of their operating wavelength. The electrical equivalent of such structure is a parallel LC circuit. This structure provides a high-impedance boundary condition for all directions of propagation and for both kinds of polarizations. They also posse in-phase reflection rather than 180° in the case of PEC. These structures are also called AMC as they imitate the boundary condition satisfied by the magnetic conductors i.e. at the surface, the tangential component of magnetic field becomes zero which is opposite to the PEC behavior where the tangential component of the electric field is zero at the conductor boundary.

In this paper, the asymmetry of the HIS cell enables it to behave differently for the incident x- and y-polarized waves. Due to slot along the x-direction, the effective length in y dimension is reduced. Hence, for the incidence of the y-polarized wave, the structure gives zero reflection at higher frequency, which can be seen in Fig. 3 [Reference Zhang, Hagen, Younis, Fischer and Wiesbeck5]. The HIS shows + reflection phase for x-polarized and for y-polarized waves at the required frequencies. This behaviour is utilized to convert the linear polarization of the antenna into circular polarization [Reference Engheta and Ziolkowski9, Reference Liang, Sanz-Izquierdo, Parker and Batchelor16].

Fig. 3.

Reflection phase (∟S11) of the unit cell (degree).

The antenna radiates a linear polarized wave that, in time-harmonic form, can be expressed as the sum of two orthogonal components in x- and y-directions. Thus, radiated electric field $(\vec{E}_d)$ from the antenna can be expressed as,

(1)$$\vec{E}_d = \; \displaystyle{{E_0} \over 2}Re\lsqb {\lpar {{\hat{a}}_x + {\hat{a}}_y} \rpar e^{\,j\lpar {\omega t-\beta z} \rpar }} \rsqb $$

The HIS gives 90° rotation to x- and − 90° rotation to y-polarized waves. Thus, the reflected electric field $(\vec{E}_r)$ is,

(2)$$\vec{E}_r = \; \displaystyle{{E_0} \over 2}Re\lsqb {\lpar {{\widehat{{\,ja}}}_x-j{\hat{a}}_y} \rpar e^{\,j\lpar {\omega t-\beta z} \rpar }} \rsqb $$

The total radiated electric field in the +z direction is superimposition of this radiated field $\vec{E}_d$ and the reflected field $\vec{E}_r$.

(3)$$\vec{E}\; = \overrightarrow {\; E_r\;} + \overrightarrow {\; E_d} = \; \displaystyle{{E_0} \over 2}Re\lsqb {\lcub {\lpar {1 + j} \rpar {\hat{a}}_x + \lpar {1-j} \rpar {\hat{a}}_y} \rcub e^{\,j\lpar {\omega t-\beta z} \rpar }} \rsqb $$

Equation (3) shows that the radiated wave has equal magnitude for x- and y-components of the waveform while being 900 out of phase, which shows that the whole assembly of the HIS and antenna radiates CP waves.

Reflection phase of HIS unit cell

The reflection phase (phase of reflection coefficient, ∟S₁₁) of the unit cell of the HIS for normally incident wave is shown in Fig. 3. Here, x-polarized waves have 90° phase at 2.5 GHz and y-polarized waves have −90° at the same frequency. The combination produces right hand circularly polarized (RHCP) waves from equation 3. Similarly, at 3.5 GHz the phase of reflected wave for the x-polarized wave is −90° and for the y-polarized wave is 90°. A left hand circularly polarized (LHCP) wave is produced from the aforementioned combination.

Reflection coefficient of bow tie antenna

The CPW fed bow tie slotted patch antenna gives resonance at frequencies of 2.5 and 3.5 GHz. The simulated S ₁₁ (reflection coefficient) for these frequencies are −43 and −35 dB respectively as shown in Fig. 4. The designed antenna shows a bandwidth of 75 MHz at both frequencies. The real WiMax bandwidth for 2.496–2.690 GHz is 5–10 MHz, and for (3.4–3.8) GHz is 5 MHz [17].

Fig. 4.

Reflection coefficient (|S11|) of the antenna.

The conventional bow-tie antenna offers a wide bandwidth because (a) the radiation depends on the angle of bow-tie opposite to the length in standard dipole antenna [Reference Balanis1]; and (b) the volume is higher which enables the more radiation modes to fit in the structure [Reference Balanis1]. The designed antenna here has a small length and width (hence less volume). Also, it is not a conventional bow-tie antenna but a bow-tie shaped slot is cut in the microstrip patch antenna. The microstrip patch antennas are inherently narrow-band structure. Hence, the designed antenna shows narrowband behavior.

Reflection coefficient and axial ratio of bow tie antenna above HIS

Figure 5 shows a bow-tie slotted patch antenna over 8 × 8 HIS aligned with 45^° shift. The separation between patch and HIS is 4.5 mm. S ₁₁ at 2.4 GHz is around −26 and −22 dB at 3.42 GHz as shown in Fig. 6. Around 100 MHz or 4% of frequency reduction is obtained at lower frequency. A total of 2.2% or 80 MHz of frequency reduction is observed at higher frequency due to placement of the HIS below antenna. Equation (3) shows the form of field for circular polarization [Reference Yi and Qu18]. The axial ratio (AR) of less than 3 dB is achieved for both bands (Fig. 7). The AR is around 2.8 dB at 2.42 GHz and 2.8 dB at 3.42 GHz.

Fig. 5.

Bow tie slotted patch antenna on top of the HIS.

Fig. 6.

Simulated and measured reflection coefficient of the antenna (solid blue line: measured, solid red line: simulations).

Fig. 7.

Simulated and measured AR of the antenna (solid red line: measured, solid black line: simulations).

Parametric study

Parametric sweep is shown in Fig. 8. As the separation between the HIS and antenna increases, the AR shifts right side and it shifts left side for a decrease in separation. The AR nearly remains constant for all sweep. The reflection coefficient with parametric sweep is observed in Fig. 9. The reflection coefficient value is shifting left and decreased as the separation between the HIS and antenna is minimized. Similar results are realized, when HIS separation is increased from 4.5 mm.

Fig. 8.

Axial ratio for different gap values.

Fig. 9.

Reflections from the antenna for different gap values.

Radiation pattern of bow tie slotted patch antenna

The radiation patterns are obtained for a bow tie slotted patch antenna at two required frequencies as shown in Figs 10 and 11. The RHCP pattern is realized in xz and yz planes while cross polarization is low in both planes at 2.4 GHz. For 3.42 GHz cross polarization is low i.e. RHCP while LHCP is high in xz and yz planes.

Fig. 10.

Radiation pattern of the antenna at 2.4 GHz.

Fig. 11.

Radiation pattern of the antenna at 3.42 GHz.

Measured reflection coefficient of proposed antenna above HIS

Figure 6 shows the simulated and measured S ₁₁ of the antenna. The measured result at higher frequency is shifted from the simulated however; at lower frequency, the result is close with a shift of 100 MHz. The measured reflection coefficient at 2.5 GHz is −27 dB and it is −20 dB for 3.5 GHz. The shift in frequencies is due to general fabrication and measurement errors.

Measured axial ratio of proposed antenna above HIS

The simulated and measured ARs are plotted in Fig. 12. The AR at lower frequency (2.5 GHz) is 2.5 dB, while for upper frequency (3.5 GHz) the AR is 0.9 dB. The measured AR shows 10 and 67% of improvement at lower and higher frequencies as compared to simulated result.

Measured radiation pattern of proposed antenna above HIS

Figures 13 and 14 show the measured radiation pattern of the antenna for the two resonating frequencies. The cross-polarization level in the measured pattern is comparable to the simulated pattern; however, more variations are obtained in the measured result. Deviation from simulated results is due to the feeding cable effect, fabricated and measurement errors.

Fig. 13.

Measured radiation pattern of the antenna at 2.4 GHz.

Fig. 14.

Measured radiation pattern of the antenna at 3.42 GHz.