II. ANTENNA CONFIGURATIONS

Figure 1 shows the configurations of the proposed antenna, which follows the principle of simplicity and practicability. It just consists of three components: a two-stage inverted cone, two shorting pins and a circular ground plane. All of the components are metal materials and connected directly. There are no unstable structures on that. The two-stage inverted cone is located at the center of the ground plane, and it can be seen as a union of a feeding cone (lower) and a radiation cone (upper). The structure of two-stage cone introduces new parameters, which can be adjusted for a better antenna performance. In practice, when the feeding cone has a smaller cone angle than that of the radiation cone, a better impedance matching and a larger bandwidth will be obtained. In addition, if the height of the entire antenna is fixed, the proportion of radiation cone and feeding cone will be a crucial parameter, which affects the impedance matching clearly in the low band. These will be discussed in the next section. The two-stage cone is integrated together and do not need to be assembled, so it is more facilitated than others [Reference Lau, Kong and Luk2, Reference Danial and Randy10, Reference Koohestani, Zurcher, Moreira and Skrivervik12]. If a light weight is required further, the cone can be manufactured into a hollow body, or the cone could be manufactured by the plastic and electroplated on the surface. The bottom of the two-stage cone connects with the inner core of the radiofrequency (RF) connector, and the circular ground plane connects with the outer shield of the RF connector. The two shorting pins are another important component. They are two thick poles and symmetrically located at two-side edges of the two-stage cone. Their effects include connecting the radiation cone to the ground plane and supporting the cone structure. In [Reference Lau, Kong and Luk2], four thin wires of 1 mm diameter are used as the grounding structure for a better performance, but they are too thin to support the radiation body. Therefore, the extra supporting structure has to be settled. In our prototype, we just use two nylon poles at the orthogonal position of the shorting pins to keep the balance of the cone (Fig. 1(c)).

Fig. 1.

Configurations of the proposed antenna. (a) Local view. (b) Full view. (c) Prototype of the antenna.

For a SWB antenna, its electrical sizes depend on the physical sizes of the antenna and the low cut-off frequency of VSWR. In our design, we determine the external dimensions of the antenna firstly and try to get the lowest cut-off frequency through the optimization of the parameters. Shown in Fig. 1, the maximum diameter of the two-stage cone 2R _p is fixed as 72 mm, and the height of the cone h _p is fixed as 26 mm. The diameter of the ground plane D _g is 296 mm, and its thickness is 2 mm. In the following section, the effect of the main parameters on the low cut-off frequency and the impedance bandwidth are analyzed. The EM simulation software CST MWS, based on Finite Integration Time Domain technology, is utilized to evaluate the performances of the wideband antenna.

III. LOW CUT-OFF FREQUENCY AND SWB PROPERTY OF THE ANTENNA

The internal parameters of the antenna mainly include those of the shorting pins and the two-stage cone. The parameter studies are carried out to investigate the antenna structure. The given sizes include: h ₁ = 12 mm, h ₂ = 10 mm, h ₃ = 3 mm, R ₁ = 14 mm, R _s  = 32.4 mm, and D _s  = 3 mm.

A) Effect of shorting pins

The cases with and without shorting pins are calculated. Figure 2 gives the VSWR curves of the two cases. The calculating frequency range is set from 0.1 to 10 GHz to identify the variation clearly in the low band. The range will be set as 0.1–20 GHz later. From Fig. 2, both cases show the high-pass characters, but the low cut-off frequency of the case with shorting pins is much lower than that without shorting pins. The former is 0.9 GHz, while the latter is merely 2.09 GHz. The low cut-off frequency is reduced by 57%. It is implied the dimensions of the antenna can be reduced more than a half by the shorting pins.

Fig. 2.

VSWR of the proposed antenna with and without shorting pins.

Figure 3 shows the input impedance curves of the two cases. Above 2 GHz, the curves of the two cases are almost same. While below 2 GHz, a new parallel resonance is aroused around 0.8 GHz after the shorting pins added. This parallel resonance enhances the values of the input resistance about 1 GHz, so the impedance matching of the cone can be solved from 0.9 to 2 GHz.

Fig. 3.

Input impedance of the proposed antenna with and without shorting pins.

B) Diameter and position of shorting pins

The parameters of the shorting pins are the diameter and the position of the pins. The VSWR curves with different diameters are shown in Fig. 4. The diameter varies from 1 to 7 mm. In a global view, the VSWR curves change a little. The main changes occur in the low band (about 1~2 GHz). After enlarging the local view of the low band, the variation of the low cut-off frequency can be clearly observed. The lowest cut-off frequency is 0.9 GHz at D _s  = 3 mm. When D _s  = 1 and 5 mm, the low cut-off frequencies are approximately equal. The frequency moves to 1.03 GHz when D _s  = 7 mm. If the shorting pins are too thin, then the extra supporting structure has to be added [Reference Lau, Kong and Luk2].

Fig. 4.

VSWR with different diameters of shorting pins.

The position of the shorting pins is expressed by a normalized parameter k _short , which equals the ratio of R _s to R _p . Figure 5 shows the VSWR curves with different k _short from 0.7 to 1. As k _short decreases, the values of VSWR in low band increase quickly and the low cut-off frequency shifts to the lower frequency slightly. When k _short  = 0.9, the low cut-off frequency and the impedance matching can achieve a good balance.

Fig. 5.

VSWR with different positions of shorting pin.

Therefore, the parameters of the shorting pins mainly affect the low cut-off frequency and the impedance matching in the low band.

C) Sizes of cone – h ₁ and R ₁

The crucial parameters of the two-stage cone are the internal ones h ₁ and R ₁, which will decide the proportion of the two stage and the cone angles. Figure 6 shows the VSWR curves with different h ₁. As h ₁ rises, the impedance matching deteriorates quickly in 1.5–3.5 GHz. When h ₁ = 9 and 12 mm, the VSWRs are both well. However, the low cut-off frequency tends toward higher and the impedance matching tends to deteriorate at 4.5 GHz when h ₁ = 9 mm.

Fig. 6.

VSWR with different h ₁.

The VSWR curves with different R ₁ are shown in Fig. 7. When R ₁ = 8 mm, the VSWR values are beyond 2 in 1.5–3.5 GHz and 6~7 GHz. As the radius increases, the impedance matching tends to balance in the whole band. When R ₁ = 20 and 26 mm, the VSWR values are beyond 2 about 4 GHz, and the low cut-off frequencies also tend toward high frequency. When R ₁ = 14 mm, the VSWR curve reaches a balance in the whole band.

Fig. 7.

VSWR with different R ₁.

In summary, h ₁ and R ₁ have little effect on the low cut-off frequency but mainly decide the impedance matching of the whole band, so they can be optimized further for the best impedance matching and bandwidth.

IV. EXPERIMENTAL RESULTS AND DISCUSSIONS

A prototype of the proposed antenna was fabricated to confirm the performances. Figure 1(c) shows the photograph of the antenna. The two-stage cone has an integrated structure. The measurement is carried out at microwave anechoic chamber at Xidian University. The frequency range of 0.1–20 GHz is observed. Figure 8 shows the measured VSWR comparing with the calculated one. The SWB property can be observed. The measured curve has the similar shape with the calculated one, but the measured values are higher than the calculated ones in medium and high band. It is due to the fabricating errors increasing as the frequency rises. The low cut-off frequencies of the two curves agree much well. The measured low cut-off frequency for VSWR ≤ 2 is 0.905 GHz, while the calculated value is 0.9 GHz. Based on 0.905 GHz, the impedance bandwidth of the proposed antenna for VSWR ≥ 2 is more than 1:22.1. The external electrical sizes of the antenna are h _p  = 0.078λ _c (26 mm) and 2R _p  = 0.217λ _c (72 mm). The diameter of the ground plane is 0.893λ _c (296 mm). All of the three electrical sizes are smaller than those of the SWB monopolar patch antenna in reference [Reference Lau, Kong and Luk2].

Fig. 8.

Measured and simulated VSWR of proposed antenna.

Such a super bandwidth almost covers the band above 1 GHz, and multi-band operations can adopt the proposed antenna. When the antenna is used in UWB systems, the time-domain properties are crucial for the antenna [Reference Gao, Yang, Niu and Zhang13]. The group delay can clearly show the non-linearity of the phase response and indicate the degree of the distortion. We use two identical proposed antennas as transmitting terminal and receiving terminal to construct a UWB link, and the distance between them is 900 mm. As the antenna is not symmetrical, two cases are calculated. One is the two antennas are placed along x-axis basing on the coordinate system of Fig. 1(b); the other is along the y-axis. Figure 9 presents the two calculated group-delay curves of the UWB system. The results show, the maximum variations of two curves are 1.7 and 1.6 ns separately, which indicates good time-domain properties of the antenna for UWB applications [Reference Gao, Hu, Cong and He14].

Fig. 9.

Group delay of the proposed antenna system.

The radiation properties of the antenna are also presented. We show the RPs of elevation plane (xoz-plane) and azimuth plane (parallel xoy-plane cutting the peak gain) at 1, 3, 9, and 20 GHz. Figure 10 collects the results. The co-polarized RPs indicate that the proposed antenna has the like-monopole omnidirectional radiation in the whole band. As the frequency increases, the peak value in elevation plane changes from high elevation angle to low one, and the co-polarized ripple level (out-of-roundness) in azimuth plane also increases. Table 1 lists the ripple levels. The maximum ripple level is not more than 6.6 dB at 20 GHz, while this level is not more than 9.9 dB in [Reference Lau, Kong and Luk2]. It implies the proposed simple structure can bring an improved roundness. Meanwhile, the cross-polarized levels are also reduced, and the worst value is about −9 dB. In comparing reference, the cross-polarized levels are almost equal to the co-polarized ones in the high-frequency band. The peak gains are also given in Table 1. As the frequency rises, the peak gain increases from 3.8 to 8.7 dB.

Fig. 10.

Measured RPs (dB). ————■————co-pol, —————cross-pol.

Table 1.

Ripple levels and peak gains against frequency.

As well known, the ground sizes of the omnidirectional antenna could affect the radiation properties, such as the elevation angle and the roundness of RPs. Here, we show the variation of the RPs at 20 GHz. Figure 11 shows the RPs of elevation plane and azimuth plane with three ground diameters, i.e. 216 mm (0.651λ _c ), 296 mm (0.893λ _c ), and 376 mm(1.134λ _c ). We can obtain the ground diameter has a little effect on the radiation properties of the proposed antenna. The elevation angle almost has no change, and the out-of-roundness are 6.4, 6.6, and 6.6 dB separately. The corresponding VSWR curves are also given in Fig. 12. The SWB property is maintained well.

Fig. 11.

RPs with different D _g at 20 GHz. (a) Elevation plane. (b) Azimuth plane.

Fig. 12.

VSWRs with different D _g .

As the summary, Table 2 shows the comparison of the proposed antenna with several literature antennas with the impedance bandwidth of more than 1:4 [Reference Lau, Kong and Luk2, Reference Danial and Randy10, Reference Koohestani, Zurcher, Moreira and Skrivervik12]. From the given results, the proposed antenna indicates the comprehensive advantages in physical structures and performances.

Table 2.

Comparison of the proposed antenna with the literature antennas.

Note: λ _c is the wavelength of the low cut-off frequency for VSWR ≤ 2.

* The antenna is square shape, so its sizes are converted into the maximal diameters.

V. CONCLUSION

A SWB and compact omnidirectional antenna is presented in the paper. The structure of the antenna is more practical and simpler than that of former SWB omnidirectional antenna, so the antenna is more suitable for the engineering. Meanwhile, the dimension of the antenna is also smaller than former SWB omnidirectional antenna. Through the simulation and the measurement, it is confirmed the antenna has the SWB impedance property and the improved like-monopole RPs. Furthermore, the roundness of the antenna in azimuth plane is improved into not more than 6.6 dB, and the cross-polarized levels are reduced by 9 dB. Based on the results, the proposed antenna has more advantages when it is used in the multi-band and UWB mobile communication systems.