Single element antenna design

The first step in constructing a BW and frequency reconfigurable MIMO antenna array is designing its unit element or antenna with a single-element radiator. A Schematic of unit element antenna is shown in Fig. 1. Cylindrical resonators are used as the primary radiator made of alumina (ceramic) material, having ${\varepsilon _r}$ = 9.8 with radius and height as 12.8 and 5 mm, respectively. Aperture coupling is used to excite the ceramic radiators. Initially, a horizontally placed rectangular aperture is designed having dimensions 17.7 × 1.75 mm². Further, two vertically oriented side slots are etched orthogonally to rectangular slot, leading to a stair-shaped aperture as displayed in Fig. 1. This unit element antenna achieves a wide impedance bandwidth from 4.79 to 5.92 GHz. To analyze the reason for achieving a wider frequency band of operation, the S₁₁ curve and electric field variations of different modification steps involved in the finalization of single-element antenna design are displayed. First of all, it can be confirmed from Fig. 3 that complete radiation is due to cylindrical resonators, as the configuration having only an aperture, excited using a line feed, is not resonating properly. Apart from this, E-field and S-parameter variations of four different configurations (Ant. 1,2,3,4) are also displayed in Fig. 3. All three antenna configurations are depicted in Fig. 2(a,b,c,d).

Figure 1.

Schematic of a single-element antenna for array design with key parameters and components.

Figure 2.

Schematic diagram of (a) Ant. 1, (b) Ant. 2, (c) Ant. 3, and (d) Ant. 4.

Figure 3.

S₁₁ and E-field distribution for all four antenna configurations.

In antenna 1, a horizontally oriented rectangular aperture is designed along with cylindrical DR and line feed. As is obvious, this structure produces HEM_11δ mode field in the cDR with (-ve) x-oriented field component, which is because the rectangular aperture acts as a magnetic dipole placed horizontally [Reference Mongia and Bhartia27].

As the effective permittivity $({\varepsilon _{eff}})$ and effective height $({H_{eff}})$ (cDR + substrate combined system) play a vital role in characterizing the behavior of the designed antenna, they can be formulated as given in [Reference Sharma, Das, Gupta and Gangwar28] using the following equations:

(1)\begin{equation}{H_{eff}} = H + {\text{ }}{H_S}\end{equation}

(2)\begin{equation}{\varepsilon _{eff}} = {\text{ }}\frac{{{H_{eff}}}}{{\frac{H}{{{\varepsilon _{rCDRA}}}} + \frac{{{H_S}}}{{{\varepsilon _{rSub}}}}}}\end{equation}

where ${\varepsilon _{rCDRA}}$, ${\varepsilon _{rSub}}$ are the relative permittivities and $H,{\text{ }}{H_S}$ represents the heights of cDR and substrate, respectively. According to [Reference Sharma, Das, Gupta and Gangwar28], the equation used for the calculation of resonant frequency for the fundamental HEM_11δ mode field is given as:

(3)\begin{align}&{f_{HE{M_{11\delta }}}} = \frac{{6.321 \times C}}{{2\pi \times R \times \sqrt {{\varepsilon _{r,eff}} + 2} }} \nonumber\\ & \left\{ {0.27 + 0.36 \left( {\frac{R}{{2 \times {H_{eff}}}}} \right) + 0.02{{\left( {\frac{R}{{2 \times {H_{eff}}}}} \right)}^2}} \right\}\end{align}

where R is radius of cDR, $C{\text{ }}$ is speed of light, ${\varepsilon _{r,eff}}$ and ${H_{eff}}{\text{ }}$ are the effective relative permittivity and effective height of antenna system.

In antenna 2, a vertical slot is added above the horizontal slot as depicted in Fig. 2(b). This vertical slot produces an E-field in the (-ve) y-direction along with the (-ve) x-oriented field of the horizontal slot. Similarly, in antenna 3, the vertical slot placed below the horizontal aperture produces a (+ve) y-directed field component. These upper and lower vertical aperture slots behave as a magnetic dipole placed vertically [Reference Kajfez, Glisson and James29]. Finally, in the 4th configuration, by combining all three aperture structures, a resultant electric field oriented at 45° angle from the x-axis is obtained, showing HEM_11δ mode fields. The individual close resonances produced by the horizontal as well as two vertical aperture slots, as visible from the S₁₁ variations in Fig. 3, are coupled together to produce a WB response. The phenomenon responsible for this behavior is “mutual resonance” or coupling resonance, which improves the impedance matching of the overall structure, generating a WB [Reference Targonski, B.Waterhouse and Pozar30].

Two-element antenna array configuration

Before designing the antenna array structure, isolation among the two antenna elements is discussed. The gap between the centers of the two radiating elements is kept at 48 mm. No power divider arrangement is used at this stage, and the two antenna elements are fed by separate ports, which are depicted in Fig. 4(a). The S-parameter curves of the aforementioned configuration are displayed in Fig. 4(b) over the operating BW. It shows that the isolation among the ports is better than 25 dB value over almost the entire frequency band [Reference Kumar and Chaudhary31]. Hence, the 50 mm gap between the two radiating elements is chosen as a reference for the proposed array geometry. In order to design the 2-element array configuration, a 3-dB power divider has been incorporated with a simplified microstrip feeding network. A λ/4 impedance transformer is utilized in the feed structure [Reference Balanis32]. Figure 5 displays the antenna array configuration. This array antenna is designed on a commercially available FR-4 epoxy substrate with an electrical relative permittivity $({\varepsilon _r}){\text{ }}$of 4.4 and loss tangent (tan δ) of 0.02. Dimensions for this configuration are chosen as 54.25 × 105 × 1.6 mm³. It is necessary to mention that DR and aperture dimensions are the same for both elements, as well as in every configuration. Figure 6(a) shows the comparative S₁₁ parameter curve of the array structure and the unit element configuration. It can be observed from it that there is a close approximation between the S₁₁ variations. Figure 6(b) shows the gain comparison between the 1 × 2 array structure as well as single radiator structure. The figure shows that the simulated gain value for the antenna array structure is nearly 2.5–3 dBi more, as compared to the single element configuration in most parts of the operating band. This happens due to the constructive interference of the radiation among the 2 elements, leading to a more directional pattern [Reference Gupta, Sharma, Das, RK and Khalily33].

Figure 4.

(a) Geometrical layout for isolation check between the two radiating elements for array design, and (b) S-parameters versus frequency curve.

Figure 5.

A 1 × 2 array geometry with 3-dB power divider line.

Figure 6.

(a) Comparative S₁₁, and (b) Gain for single radiator geometry and array design.

Figure 7.

(a) Bandpass and (b) Bandreject filter structures without diodes and biasing and their S-parameter curves, (c) Reconfigurable bandpass, (d) Bandreject filter design, (e, f) Single port reconfigurable filter + antenna array layout with biasing circuit for diodes and (g, h) S₁₁ response of structures shown in (c, d), respectively, and surface current distribution for (i) All diodes OFF, PB mode at VD cap. (j) 0.55 pF, (k) 1.18 pF, (l) 9.01 pF and NB mode at VD cap, (m) 0.72 pF, and (n) 1.18 pF.

Reconfigurable 1 × 2 array antenna configuration

This section describes the details of the frequency and BW reconfigurable two-element array antenna with three different modes of operation. In the array structure, which was already discussed in the previous section and is displayed in Fig. 5, some modifications have been incorporated within the feed network so as to adjust the lumped elements and resonating structures for frequency and BW tuning in the overall structure. This can be visualized from Fig. 7(e,f). Before integration of designed filters in the proposed array structure, their S-parameter characteristics, with and without diodes and biasing, are separately studied and is depicted in Fig. 7(a,b,g,h). The layout of reconfigurable PB and NB filters are illustrated in Fig. 7(c) and 7(d), respectively, at different capacitance values (which corresponds to a reverse bias voltage, applied in the measurement process) whereas filters without lumped elements are depicted in Fig. 7(a,b). In order to achieve a NB, a square ring-shaped resonator structure is attached in shunt to the feeding network, which behaves as an LC tank circuit. By altering the dimensions of the ring, L and C can be varied, which will further alter the resonant frequency and BW. Again, to accommodate the PB response in the overall structure, two separate dual square split ring type resonating structures are appended in shunt to the feeding network along with the NB structure. The dimensions of the two separate resonating structures are chosen to be slightly different to ensure that the notch frequencies of these resonators are also different, but not very far apart from each other. This difference is responsible for the BW of the PB response [Reference Alam, Thummaluru and Chaudhary34], which will be further exploited using VDs to achieve BW reconfigurability. All these modes are combined into the overall structure with the help of various PDs (S₁–S₁₂) and VDs (VD₁–VD₁₂). Out of the three basic modes of operations, the first one is the WB mode under which sensing could be done to further allocate some band to any particular node working for some wireless applications. In this mode, all the switches or PDs (S₁–S₆) are in OFF state. In mode 2, switches S₁, S₃, S₄, and S₆ are in ON state while S₂ and S₅ are in OFF state. This mode shows the PB response, due to the additional resonating structures connected in shunt. VD₁, VD₂, VD₄, and VD₅ being in active state, are used to tune the BW and frequency of the overall structure as VDs manipulate the dimensions of the resonating structure i.e. outer ring which shifts the frequency and also alters the BW because the inner ring resonant frequency is being fixed which varies the separation between the NB resonant frequency of the two structures.

In mode 3, switches S₂ and S₅ are ON while S₁, S₃, S₄, and S₆ are OFF. This mode shows NB response due to the square ring open-loop structure connected in shunt with the feedline, acting as an LC tank circuit. Again, the dimensions of the ring are exploited using VD₃ and VD₆, producing tunable NB [Reference Alam, Thummaluru and Chaudhary34]. For practically applying the ON/OFF states of switches, a biasing circuit is designed. A biasing voltage diagram is shown in Fig. 8(a). For brevity, the switching of only one feeding network is shown in the figure, as the diode arrangements are identical for all the feeding networks. For mode 1, as switches S₁, S₂, S₃ are all OFF, to achieve this, biasing voltages V₁ is set at 0 V and V₂, V₃, V₄ at high (5 V). The remaining can take any voltage value. For mode 2, switches S₁ and S₂ are ON and S₃ is OFF. Here, V₁ is set at 5 V, V₂, V₃ at 0 V (ground), and V₄ at a higher voltage than V₁ (10 V), while V₇ can be set at any value. Also, V₅ and V₆ control the corresponding VDs for tuning the frequency and BW. For mode 3, S₁ and S₂ are OFF, and S₃ is ON. For this to happen, V₁ is set at 5 V, V₂, V₃ at a higher voltage than V₁ (10 V), and V₄ at 0 V, whereas V₅ and V₆ can be any value. V₇ voltage controls the VD for tunability. This voltage-controlling operation is the same for each element of the array structure and finally, the MIMO configuration. Overall working of the antenna can be visualized from Table 1 shown. For mode 1, when all PDs and VDs are OFF, we get a fixed wide BW (BW 1), which can be used for spectrum sensing CR application. For mode 2, when PDs S₁, S₃, S₄, S₆ and VDs VD₁, VD₂, VD₄, VD₅ are ON, we get a tunable PB BW and frequency, going from a WB to NAB BWs (when VD reverse bias voltages are varied), utilized for communicating interweave CR operations where different application scenario needs different BWs for efficient operation. For mode 3, when PDs S₂, S₅ and VDs VD₃ and VD₆ are in ON state, we obtain tunable NB states, which rejects some portion of the spectrum (variably) and is applicable for underlay CR applications, which needs to reject the secondary user communication when it transmits above a certain threshold. Figure 7(i-n) shows the surface current distribution on feedline of proposed 1 × 2 array antenna for different modes of operation. For case 1, when all diodes are OFF, maximum current flows in the main branch of feedline and does not pass through the resonant ring structures placed in shunt with main arm. For cases 2 and 3, surface currents support tunable PB and NB response modes respectively. For this case, the maximum current bifurcates into open loop resonant structures added in parallel to main feeding arm. Further, for different capacitance values, the current value varies which manipulates the matching as well as provides shifting of resonant frequency. The optimized dimensions of the proposed structure are given as: L = 108.12, W = 105, R = 12.8, H = 5, H _S = 1.6, L ₁ = 28.5, L ₂ = 13, L ₃ = 2.1, L ₄ = 48, L ₅ = 9.5, L ₆ = 17.65, L ₇ = 8.25, L ₈ = 7.8, L ₉ = 5.25, L ₁₀ = 2.9, L ₁₁ = 2, L ₁₂ = 2.55, L ₁₃ = 1.4, L ₁₄ = 2.55, W ₁ = 3, W ₂ = 2, W ₃ = 1.25, W ₄ = 1.75, W ₅ = 1.25, W ₆ = 0.4, W ₇ = 0.2, and W ₈ = 0.5 (units in mm).

Figure 8.

(a) Diode bias voltage-controlling diagram, and (b) DC biasing circuit, PIN diodes in (c) ON state, (d) OFF state, and (e) Varactor diode equivalent circuit diagram.

Table 1.

Lumped elements (PD, VD) switching and tuning configurations for different modes of operation

Equivalent model of the diodes and biasing circuit

The RF PDs and VDs are loaded onto the feedline of the antenna in order to achieve frequency and BW tunability as well as switching among various modes of operation. The appropriate arrangement of the PDs and VDs to select different current paths is depicted in Fig. 7(f). The selected PDs and VDs are from Skyworks Inc., having model no. SMP 1345-079LF and SMV 1247-079LF, respectively. These diodes are chosen as per the frequency range of the designed antenna and work well in terms of switching and tuning performance. The equivalent model used for the PD and VD consists of a DC-blocking capacitor (C _d), a current limiting resistor (R _c), as well as RF chokes (L _R), which is displayed in Fig. 8(b). C _d blocks the biasing current flow inside the RF port, whereas L _R allows the DC flow while blocking the RF current traveling. The optimal values for C _d and L _R are 3.6 pF and 0.4 nH, respectively. The value for R _c is chosen to be 2.1 kΩ. The SMD PD used in the design works in 2 states, i.e., ON and OFF. As per the datasheet [35, 36], in the ON state of the diode, a voltage drop of 0.89 V occurs across it with 10 mA of forward current running through. Meanwhile, the diode works under reverse bias conditions in the OFF state. Practically in the ON state, PDs can be modeled as a series combination of R _ON (2.1 ohm) and L _P (0.4 nH), whereas in the OFF state, as a parallel combination of R _OFF (2.1 kohm) and C _OFF (0.15 pF) in series with L _P. The equivalent circuit for both states is depicted in Fig. 8(c,d). However, for simplicity in the simulation process, only R _ON and R _OFF are used to work in the ON and OFF states of the PDs, respectively, which could be a possible reason for the variation in S₁₁ simulated and measured outcomes for different modes. The VD is modeled as a series combination of R _S (2.1 ohm), L _S (0.4 nH), and C _J varying from 0.55 pF to 9.01 pF, which is evaluated referring to the datasheet available. The equivalent circuit is depicted in Fig. 8(e). To switch between different modes as well as tuning of frequency and BW, seven different biasing voltages are applied at various nodes denoted by V₁–V₇. PDs and VDs are controlled using these voltages.

This bias-voltage-controlling circuit is depicted in Fig. 8(a) which is already explained in the previous section.

2-Port MIMO reconfigurable antenna array configuration

After the design of a single-port 2-element RA array, it is further extended to a two-port MIMO structure in order to suppress the multipath fading effect. This could be designed basically in three ways, one is by placing the ports parallelly or anti-parallel (mirror structure), or by orienting them orthogonally. Different orientations of MIMO antenna elements provide specific advantages when implemented. Parallel orientation provides optimum spatial efficiency and enables coherent processing, orthogonal configuration obtains maximum isolation and channel independence, while antiparallel arrangement facilitates beam control, directional selectivity, as well as improved isolation [Reference Sharma, Tiwari, Singh, Kumar and Kanaujia37,Reference Nadeem and Choi38]. Hence, as per the application requirement, any of the configurations could be selected. Antiparallel orientation provides 180° phase opposition due to a mirrored arrangement while maintaining the geometric symmetry [Reference Sharma, Tiwari, Singh, Kumar and Kanaujia37]. This arrangement generates counter-propagating surface currents that naturally cancel mutual coupling components through destructive interference [Reference Nadeem and Choi38]. This configuration is also capable of generating a low envelop correlation coefficient (ECC), which is essential for effective spatial diversity [Reference Upadhyaya, Sorathiya, Al-shathri, Shafai, Patel, Pandya and Armghan39]. In this particular work, the anti-parallel configuration is selected and is displayed in Fig. 9(a,b), which is the final proposed structure. Dimensions for this MIMO structure are 108.12 mm × 105 mm × 1.6 mm³. Figure 10(a–d) shows the simulated S₁₁ parameter response for the WB mode as well as the tunability achieved in the PB and NB modes of the proposed 2-port MIMO reconfigurable array antenna. As per [Reference Petosa8], the formulation for total spectrum (TS) and tuning range (TR) is given as:

(4)\begin{equation}TS{\text{ }}\% = 2 \times \left( {\frac{{{f_{max}} - {f_{min}}}}{{{f_{max}} + {f_{min}}}}} \right) \times 100\end{equation}

(5)\begin{equation}TR{\text{ }}\% = 2 \times \left( {\frac{{{f_{high}} - {f_{low}}}}{{{f_{high}} + {f_{low}}}}} \right) \times 100\end{equation}

Figure 9.

Geometrical layout: (a) Top view and (b) Perspective view of the proposed reconfigurable 2-port MIMO antenna array structure.

Figure 10.

(a, b) Simulated S-parameters for wideband mode, (c) Partial BW and frequency reconfigurable states for PB response at different capacitance values, (d) NB frequency tunability and MIMO performance parameters, (e) ECC/DG, and (f) CCL/TARC.

Here, f_max and f_min are the maximum and minimum frequencies of operation of the proposed antenna, whereas f_high and f_low are the high and low resonance frequencies attained during the tuning.

Hence, the proposed array antenna achieves a simulated TS of 22.10% (4.83 GHz to above 6.03 GHz) and frequency TR of 18.15% (∼1.02 GHz range of tuning) for the PB response covering almost the whole wide sensing band of mode-1. The simulated BW reconfigurability is obtained with variation from 8.02% to 21.09%, with minimum BW occurring at higher capacitance values or lower applied reverse bias voltages. Figure 10(e,f) depicts the MIMO performance parameters for the proposed antenna structure. The correlation between individual antenna element’s patterns is depicted from envelope correlation coefficient (ECC) values. The overall gain of antenna system in a fading scenario is accounted for by diversity gain (DG) parameter. Using diversity techniques, these parameters can be obtained. Above mentioned MIMO parameters can be calculated as follows [Reference Sharawi40]:

(6)\begin{equation}ECC = {\text{ }}\frac{{{{\left| {S_{11}^*{\text{ }}{S_{12}} + {\text{ }}S_{21}^*{\text{ }}{S_{22}}} \right|}^2}}}{{\left( {\left( {1 - \left( {{{\left| {{S_{11}}} \right|}^2} + {\text{ }}{{\left| {{S_{21}}} \right|}^2}} \right)} \right)\left( {1 - \left( {{{\left| {{S_{22}}} \right|}^2} + {\text{ }}{{\left| {{S_{12}}} \right|}^2}} \right)} \right)} \right)}}\end{equation}

(7)\begin{equation}DG = 10{\text{ }}\sqrt {\left( {1 - {\text{ }}{{\left| {ECC} \right|}^2}} \right)} \end{equation}

To account for the maximum transmission rate that can be achieved for reliable message transfer through a channel, channel capacity loss (CCL) is calculated. CCL can be obtained using below mentioned formula [Reference Saurabh, Dubey and Meshram41]:

(8)\begin{equation}{\rho _{ii}} = 1 - {\text{ }}\mathop \sum \limits_{n = 1}^N \left| {S_n^*{S_{ni}}} \right|\end{equation}

(9)\begin{equation}{\rho _{ij}} = - {\text{ }}\mathop \sum \limits_{n = 1}^N \left| {S_n^*{S_{nj}}} \right|\end{equation}

(10)\begin{equation}{\varphi _R} = \left[ {{\rho _{ij}}} \right]\end{equation}

(11)\begin{equation}{C_{loss}} = - {\text{ }}lo{g_2}{\text{ }}det\left( {{\varphi _R}} \right)\end{equation}

Here, ${\varphi _R}{\text{ }}$denotes the correlation matrix for receive antennas, and ${\rho _{ii}},{\text{ }}{\rho _{ij}}$ depicts correlation coefficients for all i,j where i and j are not equal. In a MIMO scenario instead of reflection coefficient we use another metric called as total active reflection coefficient (TARC), which accounts for both amplitude as well as phase properties of antenna’s S-parameters. It can be calculated as [Reference Sharawi40]:

(12)\begin{equation}TARC = {\text{ }}\frac{{\sqrt {\left( {\left( {{{\left| {{S_{11}} + {\text{ }}{S_{12}}{e^{j\theta }}} \right|}^2}} \right) + {\text{ }}\left( {{{\left| {{S_{21}} + {\text{ }}{S_{22}}{e^{j\theta }}} \right|}^2}} \right)} \right)} }}{{\sqrt 2 }}\end{equation}

Figure 10(e,f) signifies better diversity performance with all parameters showing values within the standard limits in the whole operating wideband. The proposed MIMO antenna with antiparallel orientation showed good ECC values below 0.02, DG close to 10 dB, CCL (magnitude) below 0.5 bits/sec/Hz and TARC below 0 dB within the obtained bandwidth.