Antenna design and analysis

To get a fairly low RF energy from specific frequencies, a multiband antenna with increased gain is strongly recommended. On the top of the ground plane, an equiangular slot is structured. The feed line is intended to run behind the dielectric substrate. The dielectric substrate attempts to measure $80\times 80$ mm². The equiangular slotted antenna is designed on an FR4 dielectric substrate with a thickness of 1.6 mm.

The geometrical equilateral triangular slot has a side length (L) of 54 mm. A 50 Ω resistive microstrip transmission line with an optimum distance (L_f) of 66 mm and a width (W_f) of 3.1 mm energises this suggested slot. According to [Reference Chang and Kiang44], the idea of variable FL has been considered. By keeping the slot dimension unchanged, the length of the feed line is only varied. Figure 1 illustrates the layout and construction designs of the spatial design slot antenna without reflector. The requisite computer models for the presented antenna’s assessment were decided to be carried out with the comprehensive electromagnetic simulation platform CST Studio Suite. Before designing the proposed slot on a simulation tool, the important dimensional sizes are determined by calculation.

Figure 1.

The proposed triple-band fractal slot antenna without reflector has a layout and construction prototypes.

Simulation and measurement results

After the successful completion of all requisite simulation studies, this same proposed antenna is crafted on the S103 LPKF ProtoMat machine. For linking to the RF cable, the antenna is connected to a 50 Ω inbuilt impedance SubMiniature Version A (SMA) connector. The E8363C Power Network Analyzer (PNA) network analyzer is used for measuring the antenna’s input reflection coefficient initially. Figure 2 demonstrates the variation of $|S_{11}|$ versus resonant frequency of the designed antenna without the need for a reflective surface.

Figure 2.

Measurement of the proposed antenna’s coefficient of reflection ($|S_{11}|$) frequency dependence.

Simulation and measurement results are plotted in a single graph for comparison. Figure 3 represents the computation and recorded outcomes of the proposed antenna’s reflection coefficient ($|S_{11}|$) versus frequency. In accordance with the numerical simulations, the antenna resonates at 2.6, 3.5, and 4.9 GHz. The measuring device results were in close agreement with the numerical simulations. The characteristic impedance of the measured result ($|S_{11}|$) is less than −15 dB for all three bands. Figure 4 shows the parameterized investigation of the proposed antenna’s FL versus switching frequency. The achievement of $|S_{11}|$ is analyzed by expanding the FL by 10 mm every time from 30 to 70 mm. At an FL of 66 mm, it is feasible to obtain consistent triple bands of WLAN and WiMAX frequencies. The simulation findings reveal that for the three control RFs, the characteristic impedance was lower than −25 dB. Figure 5 illustrates the computation outcomes of the proposed antenna’s three-dimensional beams without the need for a reflective surface. The radiation patterns show that at the resonant frequencies of 2.6, 3.5, and 4.9 GHz the realized gain of more than 3 dB has been achieved. The above patterns are bi-directional and directed along the +Z and −Z axes. The gain of the antenna has been enhanced while attempting to maintain it above the operating frequencies by employing a periodic array of a planar structure. The following section briefly describes the architectural features, construction, and measurement of a symmetrical DRS planar array, along with the proposed antenna.

Figure 3.

The magnitude of the coefficient of reflection ($|S_{11}|$) frequency dependence of the antenna without the need for a reflector was simulated and measured.

Figure 4.

Parametric analysis of variation of feed length (FL) without reflector.

Figure 5.

The computational results of the proposed antenna’s three-dimensional radiation patterns without a reflector.

Defected reflector structure

Figure 6 depicts the suggested planar DRS layout and construction models. This reflector structure is modeled after the reflector introduced in [Reference Aboualalaa, Rahman, Allam, Elsadek and Pokharel4], with the objective of increasing the antenna’s gain at three operating frequencies without attempting to change their resonances. The size of the reflective surface is $80 \times 80$ mm², which really is equivalent to the size of the antenna. Square rings of a $7\times 7$ array are spaced 1 mm apart within this reflector. The square ring’s side length is 10 mm and its thickness is 0.5 mm. The distance between the last square ring and the substrate end is 2 mm. These square rings are designed periodically on the substrate, with no metal printed on the bottom side. The presented reflector is constructed on a substrate surface (FR4) with properties similar to those used in antenna design.

Figure 6.

Schematic and fabrication prototypes of the proposed planar defected reflector structure (DRS).

Results of the antenna with DRS

Figure 7 characterizes computation and construction configuration settings of the antenna with DRS. Teflon sticks with a diameter of 8 mm are employed to facilitate the antenna, which would be kept separate from the reflector. The distance between the antenna and the reflector is streamlined to be 0.54$\lambda/4$ (approximately 15 mm). Figure 8 illustrates a picture of the coefficient of reflection ($|S_{11}|$) function of frequency of the antenna with DRS. The network analyzer’s display is also enlarged for viewing the triple resonant bands. Figure 9 shows that the antenna’s simulation results with reflector distance (d). The simulation model for the antenna with the reflective surface is again managed to be performed by vastly differing the reflector distance of 5 mm each time from d = 5 mm to d = 20 mm. At a reflector length of d = 15 mm, the directional antennas’ continuous resonance frequency tunes of 2.7, 3.5, and 4.8 GHz related frequency bands are acquired. Figure 10 conveys the modeling and evaluated findings for the proposed directional antennas’ coefficient of reflection ($|S_{11}|$) frequency dependence. The first two bands are well suited, whereas the third frequency range has a small gap with the simulated results due to impedance mismatch in tolerance levels.

Figure 7.

Computational and experimental configurations of the antenna with DRS reflector.

Figure 8.

The magnitude of the coefficient of reflection ($|S_{11}|$) frequency dependence of the proposed antenna with DRS reflective surface was measured.

Figure 9.

The parameterized evaluation of the antenna’s simulated data with distance (d).

Figure 10.

The magnitude of the coefficient of reflection ($|S_{11}|$) with DRS reflective surface was designed to simulate and analyzed.

The defected reflector structure (DRS) is suitable to place behind the antenna at a small distance to improve the gain of the proposed antenna. Figure 11 portrays the modeling outcomes of the proposed antenna’s three-dimensional radiation patterns with DRS at three frequency bands. Gain values of more than 3 dB were achieved without attempting to change the resonance frequency. DRS achieved the enhanced gains of 6.42, 9.26, and 8.1 dB at frequencies of 2.7, 3.5, and 4.8 GHz, respectively. The main beam of unidirectional radiation patterns for the three frequencies could really be observed to be directly aimed toward the +Z-direction. A transmit antenna with a broadband gain is employed for gain and radiation pattern measurement techniques. The antenna is positioned at the receiving end, at a distance of 1.4 m from the transmit antenna (horn). To measure the antenna gain and radiation patterns, network analyzer port-1 is attached to the transmit antenna and port-2 is linked to the receiver side (proposed). The radiation patterns at every one of the three operating frequencies are measured and displayed along with the the numerical simulations. With DRS, the proposed antenna measured gain values of 6.1, 8.8, and 7.6 dB at harmonic resonance of 2.7, 3.5, and 4.8 GHz, respectively. The field radiation patterns are standardized to zero, which would be comparable toward how numerical simulations are linearly proportional in the CST simulation software. The modeling and measurement data of relatively stable patterns of the suggested antenna with DRS at mainly two planes of XZ-plane ($\phi=0^{\circ}$) and YZ-plane ($\phi=90^{\circ}$) patterns are displayed in Fig. 12. It should be acknowledged that the this double patterns are uni-directive and focused along the +Z-axis. To enhance gain and radiation effectiveness, a metallic reflector is usually situated at a distance of $\lambda/4$ behind the antenna. The DRS is positioned behind the antenna at a length of 0.54$\lambda/4$ in the prototype architecture. The designed antenna has the benefits of a compact design, simple fabrication, and connection adaptability to the rectifier.

Figure 11.

Radiation patterns with DRS reflector at three operating frequencies were simulated.

Figure 12.

Normalized radiation patterns with DRS reflector.

The proposed antenna’s equivalent circuit is shown in Fig. 13. The parallel RLC circuit and inductor in series constitute the microstrip feed line. The fractal slot and its coupling with the microstrip feed line are represented by the two parallel LC circuits that are both in series with capacitors. Simulation and measured results of gain of the proposed antenna is shown in Fig. 14. Experimental results are well in agreement with the simulation results.

Figure 13.

The proposed antenna equivalent circuit.

Figure 14.

The gain of the proposed antenna (simulation and measurement results).

Rectifier design and analysis

For converting collected high-frequency microwave power into DC, a microstrip transmission line-based triple-band rectifier with an impedance matching network is proposed. Figure 15 shows the design diagram of the rectifier. Table 1 shows the geometrical attributes of microstrip lines utilized during the layout of the rectifier. The capacitor C ₁ serves as a DC block capacitor, shielding the RF source from unwanted harmonics generated by a diode’s nonlinear device [45].

Figure 15.

Design diagram of the rectifier circuit.

Similarly, the capacitor C ₂ is useful for passing only direct current power to the resistive load. [Diode model-SMS7630-079LF (D ₁, D ₂) from Skyworks and capacitors $C_{1}=C_{2}=100$ pF]. In nature, RF signals exhibit sinusoidal behavior, with continuous periodic waves of equal positive and negative amplitudes. The proposed rectifier is built on 1.52-mm-thick Rogers RO4003C substrate. The simulation model needed for the evaluation of the suggested rectifier are managed to perform using the relatively high frequency simulator mostly in advanced design system (ADS). Figure 16 shows the proposed triple-band rectifier’s layout configuration setup. The modeling and evaluated outcomes of the rectifier are discussed in the subsequent section.

Figure 16.

Layout configuration of the rectifier.

Results and discussion

After the completion of the necessary simulation studies, the rectifier was manufactured on the PCB machine. The rectifier circuit includes the necessary components of a capacitor, diode, and optimum resistive load for measuring the reflection coefficient and output voltage. The rectifier is now connected to the analyzer (which also provides an RF signal generator) through an SMA connector with an internal impedance of 50 Ω. First, we were using the E8363C PNA network analyzer for measuring the reflection coefficient ($|S_{11}|$) plotted as a function of the suggested rectifying circuit and especially in comparison to the simulated results. Figure 17 demonstrates the computation and evaluated results for the rectifier coefficient of reflection frequency dependence. In the ADS simulator, the S-parameter modeling is used to acquire the triple bands of the proposed rectifier. The first and third bands of the evaluated coefficient of reflection resonance frequency match the simulated results, whereas the second band has a small gap due to impedance mismatch between the input impedance and the rectifier circuit. The proposed rectifier is constructed and simulated for the required frequency bands of 2.7, 3.5, and 4.8 GHz, which seem to be nearer to the antenna resonance frequencies.

Figure 17.

Reflection coefficient ($|S_{11}|$) of the rectifier (input power = 0 dBm, resistive load = 1 kΩ).

Table 1.

Dimensional values of rectifier circuit

The suggested rectifier’s frequency response has a magnitude of coefficient of reflection $|S_{11}|$ less than −15 dB. The simulation model in the ADS simulator is decided to continue to approximate the output voltage at different loading resistance values and input powers utilizing harmonic balance modeling. Modifying the input power from −15 to 20 dBm at specific load resistance values that range from 0.5 to 2.5 kΩ produces output voltages. Figure 18 illustrates the modeling outcomes for rectifier’s output voltage. The lower output voltage is obtained at 0.5 kΩ, whereas the higher output voltage is obtained at 2.5 kΩ. The output voltage changes significantly when the input power is changed from −15 to 5 dBm. When the input power exceeds 5 dBm, however, the output voltage becomes saturated. The maximum modeling voltage output of 2.18 V is acquired by concurrently applying all the three frequency responses at the input power of 20 dBm and a resistive load of 2.5 kΩ. Using modeling output voltage values and their corresponding powers and resistive load, the efficiency is determined by calculating equation (1) [Reference Liu, Zhang and Yang15].

Figure 18.

Rectifier’s output voltage with input power at various resistive loads.

(1)\begin{equation} \text{Efficiency}\ (\%) = \frac {V_{\textrm {0}}^{2} /R_{\textrm {L}}}{P_{\textrm {in}}}\times 100. \end{equation}

Figure 19 illustrates the modeling outcomes of the proposed rectifier’s determined efficiency versus input power at different loading resistances. At a power input of 0 dBm and a resistive load of 1 kΩ, the suggested rectifier reached a maximum simulation model conversion efficiency of 79.8%. Because of the very small increment of output voltages obtained at these input powers, the conversion efficiency gradually decreases beyond the input power of 0 and 5 dBm. Except for the load impedance of 0.5 kΩ, the suggested rectifier does have an efficiency of more than 30% at relatively low powers of less than 0 dBm for the load resistance values noted in the modeling. As a result, the presented triple-band rectifier is considered suitable for attempting to convert relatively low RF powers into DC. Following the completion of the simulation studies, the rectifier’s output voltage is determined utilizing joint optimization of 1 kΩ load impedance and 0 dBm power supplied. The SMA port of the rectifier is linked to a network analyzer (which serves as an RF power source), and the voltmeter is connected to the load resistance for output voltage measurement. Figure 20 illustrates an experimental set up for measuring the DC voltage output of the proposed rectifier. The measurement photograph shows that the digital multimeter has a DC voltage output of 1.127 V at the input RF power of 5 dBm. Figure 21 represents the computation and measured output voltage and effectiveness of the rectifier at structure and functionality of 1 kΩ resistive load (R _L) and 0 dBm input power (P _in). At the input power of 0 dBm, the peak output voltage and rectifier’s efficiency obtained are 1.78 V and 74.3%, respectively.

Figure 19.

Calculated efficiency with input power at various resistive loads.

Figure 20.

The proposed rectifier’s experimental setup for taking measurements of output DC voltage.

Figure 21.

The proposed rectifier’s output voltage and efficiency were designed to simulate and evaluated at optimized values.