Structure of antenna

The structure of the proposed fully symmetric dual-polarized dual-port antenna based on differential feeding is shown in Figure 1, which consists of two layers of FR4 dielectric substrates (ε_r = 4.4, tanδ = 0.02) and three metal layers. The top radiating structure is a rectangular patch (Patch1) with a slot, which is located at the junction between the green patch and the red cross-structure patch with a slot width S of 0.1 mm. The middle metal layer is a circular patch (Patch2), which is used for improving the impedance matching characteristics of the antenna at high frequencies. The ground plane is on the bottom. Four metallized vias function as coaxial feeding probes, constituting two pairs of differential ports: P_{d 1} (P1/P3) and P_{d 2} (P2/P4).

Figure 1.

Structure of the proposed antenna: (a) perspective view, (b) Patch1, (c) front view, and (d) top view of the middle metal layer (size parameters: H1 = 3.2, H2 = 3.2, Wp = 12.8, Lst = 6.4, Wst = 3, S = 0.1, Rsub = 22, Rp = 2.8, Rf = 3.9, Rny = 19, Sr1 = 0.3, and Sr2 = 0.5 [unit: mm]).

Design of radiating patch

First, the differential S-parameters can be calculated by [Reference Eisenstadt, Stengel and Thompson27].

(1)\begin{equation}{S_{dd11}} = ({S_{11}} - {S_{12}} - {S_{21}} + {S_{22}})/2\end{equation}

(2)\begin{equation}{S_{dd22}} = ({S_{33}} - {S_{34}} - {S_{43}} + {S_{44}})/2\end{equation}

(3)\begin{equation}{S_{dd21}} = \left( {{S_{31}} - {S_{41}} - {S_{32}} + {S_{42}}} \right)/2\end{equation}

(4)\begin{equation}{S_{dd12}} = \left( {{S_{13}} - {S_{14}} - {S_{23}} + {S_{24}}} \right)/2\end{equation}

Since the antenna is of a fully symmetric structure, S ₃₁ = S ₃₂ and S ₄₁ = S ₄₂, and their coupling coefficients (S_{dd 21} and S_{dd 12}) are theoretically zero [Reference Ta and Nguyen-Trong28]. Next, we will use the defined S-parameters to evaluate the performance of antenna structural evolution.

Figure 2 illustrates the structural evolution from the original structure (Ant. 1) to the proposed antenna. The initial design (Ant. 1) uses a traditional square patch (green area) combined with four small rectangular strips (pink area) separated by white slots whose width is S. Each small rectangular patch connects to a coaxial feed probe (marked by red circles), forming the Γ-probe. In Figure 2, a side view of the Γ-probe is shown above Ant. 1, where the red cylinder represents the coaxial feed probe, and the upper pink rectangular patch is the strip. Figure 3(a) shows the |S_{dd 11}| of Ant. 1 (with Γ-probe) and a typical square patch antenna with four differential probes (without Γ-probe). It can be seen that Ant. 1 can generate two resonant frequencies f ₁ and f ₃ and the Γ-probe design can significantly widen the bandwidth. According to the current distributions for f ₁ and f ₃ shown in Figure 3(b) and (c), at f ₁, the dominant current is in-phase, exhibiting the TM₁₀ mode. At f ₃, the current distribution is even-symmetric along the x-axis and odd-symmetric along the y-axis, corresponding to the TM₁₂ mode.

Figure 2.

Design evolution of the proposed antenna configuration.

Figure 3.

(a) |S_{dd 11}| of Ant. 1, (b) surface current at f ₁, and (c) surface current at f ₃.

Next, by extending the four Γ-probe-based strips into a pink cross structure, Ant. 2 is developed. This structure can introduce a new resonant point f ₂ at 5.8 GHz, as shown in Figure 4(a). f ₂ is directly affected by the length of the cross strips Lst, which means f ₂ can shift to a lower frequency when Lst increases. At f ₂, Figure 4(b) shows that the current is primarily concentrated within the interior and edges of the cross-shaped structure. The dominant current is in-phase, indicating that the TM₁₀ mode is generated by the cross-shaped structure.

Figure 4.

(a) |S_{dd 11}| of Ant. 2 under different Lst and (b) surface current at f ₂.

We further extend the cross strips of Ant. 2 outward until they can truncate the green patch and Ant. 3 is created. This new structure can excite one more slot mode at f ₄ (with current concentrated around the slot), as shown in Figure 5(a). Figure 5(a) also indicates that adjusting Wst (i.e., changing the position and length of the slot) can independently control the frequency of f ₄. In addition, to solve the impedance mismatch of Ant. 3 at f ₁, a circular slot ring (marked by the white circle) is etched at the feeding point to introduce a compensating capacitance, counteracting the inductive effect of the probe [Reference Hall29]. At this stage, the antenna (with an etched slot at the feeding probe) has a good impedance matching property at f ₁ but that at high frequency f ₄ needs to be optimized. f ₄ corresponds to a slot mode generated by the gap between the patches. To optimize the impedance matching at f ₄, a blue circular patch (Patch2) is added at the height H1 beneath the center of the patch to modify the magnetic and electric fields of the slot, thereby improving the impedance matching at f ₄. This modification leads to the final design (Prop.). The performance of the two antennas is shown in Figure 6(a). We can clearly see that the operating bandwidth of Prop. antenna is effectively expanded by adding Patch2 to 71.5% (4.06–8.58 GHz). Additionally, the calculation for the input impedance of the differential-fed patch antenna can be expressed as [Reference Zhang and Wang30]. It is evident from Figure 6(b) that the impedance matching has been significantly improved after the circular slot ring is added.

Figure 5.

(a) |S_{dd 11}| of Ant. 3 under different Wst and (b) surface current at f ₄.

Figure 6.

(a) |S_{dd 11}| and (b) input impedance of Ant. 3, Ant. 3 with etched slot and Prop.

FR4’s high loss tangent can degrade the antenna’s efficiency. According to the previous studies, using a thicker FR4 substrate can mitigate this effect [Reference Nascimento, Lacava and Microstrip Antennas31, Reference Nascimento and Lacava32]. A low-cost probe-fed microstrip antenna using a 6.6 mm FR4 substrate can achieve nearly 75% radiation efficiency [Reference Nascimento, Lacava and Microstrip Antennas31]. However, these designs use a single-probe feed, which raises cross-polarization and adds inductive reactance. Efficiency is improved by adopting a dual-probe feed [Reference Petosa, Ittipibon and Gagnon33] and etching a capacitive ring slot near the probe(s) to compensate for the inductive reactance [Reference Hall29, Reference Nascimento, Mores, Schildberg and Lacava34], as shown in Figure 7.

Figure 7.

Comparison of total efficiency for four different antenna configurations: single-probe feed, dual-probe feed, single-probe feed with etched capacitive ring slot, and the proposed design (dual-probe feed with etched capacitive ring slot).

Realization of differential feed

To meet the far-field radiation pattern testing requirements for broadband antennas, we adopt the 180° wideband power divider (WPD) [Reference Bialkowski and Abbosh35–Reference Ta, Nguyen, Nguyen, Ngoc and Trong37] as the differential feeding, as shown in Figure 8(a). It has a T-type architecture that combines slot line and microstrip transmission line, with an integrated microstrip-to-slot line broadband transition network. The simulated S-parameters of the designed WPD are shown in Figure 8(b), with |S ₁₁| < −10 dB ranging from 3.5 to 8.5 GHz and balanced power distribution at the output ports (|S ₂₁| = |S ₃₁| = −3.9 ± 0.4 dB). In addition, the differential output ports (Port 2/Port 3) demonstrate a consistent 180° phase difference with phase variance within ±6° across the operational bandwidth.

Figure 8.

(a) Geometry of the 180° WPD (unit: mm) and (b) simulated (Sim.) and measured (Mea.) S-parameters of the 180° WPD.

Measurements

To validate the proposed wideband dual-polarized microstrip antenna, we fabricated a prototype of the antenna, as shown in Figure 9(a). Nylon screws are used to fix the two-layer structure, and the RG402 coaxial cables are soldered to the four ports (P1–P4). The overall size of the antenna is 44 × 44 × 6.4 mm³. First, the S-parameters of the four antenna ports are measured. The differential S-parameters are then calculated using formulas (1), (2), (3), and (4) based on the measured S-parameters of the four ports. The simulated and measured differential S-parameters are presented in Figure 9(b). The simulated and measured results show good agreement, with the measured impedance bandwidth (|S₁₁| < −10 dB) reaching 71.5% (4.06–8.58 GHz) while port isolation remains above 42 dB throughout the operating band. Under Eq. (3) and Eq. (4), when the two cable lines connecting to the differential port are perfectly 180° out of phase, S_{dd 21} and S_{dd 12} are theoretically zero [Reference Ta and Nguyen-Trong28]. Thus, the inter-port isolation mainly depends on the phase matching of the cables between the antenna and the differential network.

Figure 9.

(a) Antenna prototype and (b) simulated (Sim.) and measured (Mea.) S-parameters of antenna.

Next, similar to [Reference Wu, Guo, Zhu, Jiang and Ye36, Reference Ta, Nguyen, Nguyen, Ngoc and Trong37], the differential ports (Port 2/Port 3) of the WPD are connected to the antenna’s differential ports P_{d 1} (P1/P3) or P_{d 2} (P2/P4), as shown in Figure 10(a). After passing through the WPD, the signal is split into two differential signals that are directed to (Port 2/Port 3). The far-field radiation, gain and total efficiency of the antenna are then measured in a microwave anechoic chamber. The gain and total efficiency of the antenna within the operating frequency range are shown in Figure 10(b). The measured gain ranges from 3.5 to 8.2 dBi in the operating frequency band of 4.06–8.58 GHz, slightly lower than the simulation values. This small discrepancy may be caused by the fabrication and measurement errors.

Figure 10.

(a) Far-field measurement environment and the antenna with feeding circuits and (b) realized gains and total efficiency of antenna.

Figures 11 and 12 show the radiation patterns of the antenna under the excitations at differential ports P_{d 1} and P_{d 2}, respectively. By using the differential feeding scheme, the simulated cross-polarization levels (<−30 dB) are very low. Due to the presence of an asymmetric power divider, the measured cross-polarization levels are slightly higher than the simulated ones but they remain less than −20 dB across the operating frequency band.

Figure 11.

The normalized radiation pattern of the antenna under excitation at differential port P_{d 1}.

Figure 12.

The normalized radiation pattern of the antenna under excitation at differential port P_{d 2}.

Comparison and discussion

Table 1 compares the key technical parameters of the differential-fed dual-polarized antenna with some related researches. All of these designs can achieve high isolation between 35 and 52.5 dB by differential feeding and symmetric structures. Compared to other studies, the antenna proposed in this paper has a low profile with only 0.086λ_L, high isolation of 42 dB and broad bandwidth of 71.5% with relatively stable radiation characteristics, providing a promising solution for practical engineering IBFD applications.

Table 1.

Comparison of the differential-fed dual-polarized antennas with low-profile and broadband characteristics